Antenna structures and methods thereof for determining a frequency offset based on a reactance measurement

ABSTRACT

A system that incorporates the subject disclosure may include, for example, a circuit for measuring a change in reactance of an antenna, determining a frequency offset of the antenna based on a change in an operating frequency of the antenna according to the change in reactance of the antenna, and adjusting the operating frequency of the antenna to mitigate the frequency offset of the antenna. Other embodiments are disclosed.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application claims the benefit of priority to U.S.Provisional Application No. 61/896,233 filed on Oct. 28, 2013, which ishereby incorporated herein by reference in its entirety.

The present application claims the benefit of priority to U.S.Provisional Application No. 61/932,831 filed on Jan. 29, 2014, which ishereby incorporated herein by reference in its entirety.

The present application claims the benefit of priority to U.S.Provisional Application No. 61/941,888 filed on Feb. 19, 2014, which ishereby incorporated herein by reference in its entirety.

FIELD OF THE DISCLOSURE

The present disclosure relates generally to antenna structures andmethods thereof for determining a frequency offset based on a reactancemeasurement.

BACKGROUND

It is common for communications devices to have multiple antennas thatare packaged close together (e.g., less than a quarter of a wavelengthapart) and that can operate simultaneously within the same frequencyband. Common examples of such communications devices include portablecommunications products such as cellular handsets, personal digitalassistants (PDAs), and wireless networking devices or data cards forpersonal computers (PCs). Many system architectures (such as MultipleInput Multiple Output (MIMO)) and standard protocols for mobile wirelesscommunications devices (such as 802.11n for wireless LAN, and 3G and 4Gdata communications such as 802.16e (WiMAX), HSDPA, 1xEVDO, and LTE) mayrequire multiple antennas operating simultaneously.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference will now be made to the accompanying drawings, which are notnecessarily drawn to scale, and wherein:

FIG. 1A illustrates an antenna structure with two parallel dipoles;

FIG. 1B illustrates current flow resulting from excitation of one dipolein the antenna structure of FIG. 1A;

FIG. 1C illustrates a model corresponding to the antenna structure ofFIG. 1A;

FIG. 1D is a graph illustrating scattering parameters for the FIG. 1Cantenna structure;

FIG. 1E is a graph illustrating the current ratios for the FIG. 1Cantenna structure;

FIG. 1F is a graph illustrating gain patterns for the FIG. 1C antennastructure;

FIG. 1G is a graph illustrating envelope correlation for the FIG. 1Cantenna structure;

FIG. 2A illustrates an antenna structure with two parallel dipolesconnected by connecting elements in accordance with one or moreembodiments of the disclosure;

FIG. 2B illustrates a model corresponding to the antenna structure ofFIG. 2A;

FIG. 2C is a graph illustrating scattering parameters for the FIG. 2Bantenna structure;

FIG. 2D is a graph illustrating scattering parameters for the FIG. 2Bantenna structure with lumped element impedance matching at both ports;

FIG. 2E is a graph illustrating the current ratios for the FIG. 2Bantenna structure;

FIG. 2F is a graph illustrating gain patterns for the FIG. 2B antennastructure;

FIG. 2G is a graph illustrating envelope correlation for the FIG. 2Bantenna structure;

FIG. 3A illustrates an antenna structure with two parallel dipolesconnected by meandered connecting elements in accordance with one ormore embodiments of the disclosure;

FIG. 3B is a graph showing scattering parameters for the FIG. 3A antennastructure;

FIG. 3C is a graph illustrating current ratios for the FIG. 3A antennastructure;

FIG. 3D is a graph illustrating gain patterns for the FIG. 3A antennastructure;

FIG. 3E is a graph illustrating envelope correlation for the FIG. 3Aantenna structure;

FIG. 4 illustrates an antenna structure with a ground or counterpoise inaccordance with one or more embodiments of the disclosure;

FIG. 5 illustrates a balanced antenna structure in accordance with oneor more embodiments of the disclosure;

FIG. 6A illustrates an antenna structure in accordance with one or moreembodiments of the disclosure;

FIG. 6B is a graph showing scattering parameters for the FIG. 6A antennastructure for a particular dipole width dimension;

FIG. 6C is a graph showing scattering parameters for the FIG. 6A antennastructure for another dipole width dimension;

FIG. 7 illustrates an antenna structure fabricated on a printed circuitboard in accordance with one or more embodiments of the disclosure;

FIG. 8A illustrates an antenna structure having dual resonance inaccordance with one or more embodiments of the disclosure;

FIG. 8B is a graph illustrating scattering parameters for the FIG. 8Aantenna structure;

FIG. 9 illustrates a tunable antenna structure in accordance with one ormore embodiments of the disclosure;

FIGS. 10A and 10B illustrate antenna structures having connectingelements positioned at different locations along the length of theantenna elements in accordance with one or more embodiments of thedisclosure;

FIGS. 10C and 10D are graphs illustrating scattering parameters for theFIGS. 10A and 10B antenna structures, respectively;

FIG. 11 illustrates an antenna structure including connecting elementshaving switches in accordance with one or more embodiments of thedisclosure;

FIG. 12 illustrates an antenna structure having a connecting elementwith a filter coupled thereto in accordance with one or more embodimentsof the disclosure;

FIG. 13 illustrates an antenna structure having two connecting elementswith filters coupled thereto in accordance with one or more embodimentsof the disclosure;

FIG. 14 illustrates an antenna structure having a tunable connectingelement in accordance with one or more embodiments of the disclosure;

FIG. 15 illustrates an antenna structure mounted on a PCB assembly inaccordance with one or more embodiments of the disclosure;

FIG. 16 illustrates another antenna structure mounted on a PCB assemblyin accordance with one or more embodiments of the disclosure;

FIG. 17 illustrates an alternate antenna structure that can be mountedon a PCB assembly in accordance with one or more embodiments of thedisclosure;

FIG. 18A illustrates a three mode antenna structure in accordance withone or more embodiments of the disclosure;

FIG. 18B is a graph illustrating the gain patterns for the FIG. 18Aantenna structure;

FIG. 19 illustrates an antenna and power amplifier combiner applicationfor an antenna structure in accordance with one or more embodiments ofthe disclosure;

FIGS. 20A and 20B illustrate a multimode antenna structure useable,e.g., in a WiMAX USB or ExpressCard/34 device in accordance with one ormore further embodiments of the subject disclosure.

FIG. 20C illustrates a test assembly used to measure the performance ofthe antenna of FIGS. 20A and 20B.

FIGS. 20D to 20J illustrate test measurement results for the antenna ofFIGS. 20A and 20B.

FIGS. 21A and 21B illustrate a multimode antenna structure useable,e.g., in a WiMAX USB dongle in accordance with one or more alternateembodiments of the subject disclosure.

FIGS. 22A and 22B illustrate a multimode antenna structure useable,e.g., in a WiMAX USB dongle in accordance with one or more alternateembodiments of the subject disclosure.

FIG. 23A illustrates a test assembly used to measure the performance ofthe antenna of FIGS. 21A and 21B.

FIGS. 23B to 23K illustrate test measurement results for the antenna ofFIGS. 21A and 21B.

FIG. 24 is a schematic block diagram of an antenna structure with a beamsteering mechanism in accordance with one or more embodiments of thesubject disclosure.

FIGS. 25A, 25B, 25C, 25D, 25E, 25F, and 25G illustrate a comparison ofantenna pattern test measurement results.

FIG. 26 illustrates the gain advantage of an antenna structure inaccordance with one or more embodiments of the subject disclosure as afunction of the phase angle difference between feedpoints.

FIG. 27A is a schematic diagram illustrating a simple dual-band branchline monopole antenna structure.

FIG. 27B illustrates current distribution in the FIG. 27A antennastructure.

FIG. 27C is a schematic diagram illustrating a spurline band stopfilter.

FIGS. 27D and 27E are test results illustrating frequency rejection inthe FIG. 27A antenna structure.

FIG. 28 is a schematic diagram illustrating an antenna structure with aband-rejection slot in accordance with one or more embodiments of thesubject disclosure.

FIG. 29A illustrates an alternate antenna structure with aband-rejection slot in accordance with one or more embodiments of thesubject disclosure.

FIGS. 29B and 29C illustrate test measurement results for the FIG. 29Aantenna structure.

FIG. 30 depicts an illustrative embodiment of an antenna structure inaccordance with one or more embodiments;

FIG. 31 depicts an illustrative embodiment of a multiband antennastructure in accordance with one or more embodiments;

FIGS. 32A, 32B and 32C illustrate tuning using discrete selection ofinductance to select antenna fundamental resonance frequency inaccordance with one or more embodiments;

FIGS. 33A, 33B and 33C illustrate tuning using discrete selection ofinductance to select fundamental resonance frequency where a separatebut co-located high band element is shown with feed points F1H and F2Hthat allows for compatibility with RF transceiver front end designsrequiring separate low- and mid- or low- and high- band connections tothe antenna in accordance with one or more embodiments;

FIGS. 34A, 34B and 34C illustrate tuning and filtering using discreteselection of inductance to select antenna fundamental resonancefrequency in accordance with one or more embodiments;

FIGS. 35A, 35B and 35C illustrate tuning and filtering using discreteselection of inductance to select fundamental resonance frequency inaccordance with one or more embodiments;

FIGS. 36A-36B depict illustrative embodiments of a near field sensor;

FIG. 37 depicts illustrative embodiments for placement of the near fieldsensor as a probe;

FIG. 38 depicts illustrative embodiments of environmental use casesapplied to the antenna structure of FIG. 27;

FIG. 39 depicts illustrative embodiments of return loss and efficiencyplots according to the environmental use cases of FIG. 38;

FIG. 40 depicts illustrative embodiments of Smith charts according tothe environmental use cases of FIG. 38;

FIG. 41 depicts illustrative embodiments of magnitude and phase plotsassociated with one of the probes;

FIG. 42 depicts illustrative embodiments of magnitude and phase plotsassociated with one of the probes;

FIG. 43 depicts illustrative embodiments of phase vs. antenna frequencyshift plots of the probes;

FIG. 44 depicts illustrative embodiments of probe power shift vs.antenna radiated power shift plots of the probes;

FIG. 45 depicts illustrative embodiments of an antenna structure andprobe placements;

FIG. 46 depicts illustrative embodiments of a free space resonancetuning use a Ltune (a tuning variable);

FIG. 47 depicts illustrative embodiments of efficiency gain by tuning ofthe antenna structure based on measurements of one of the probes of theantenna structure of FIG. 45;

FIG. 48 depicts illustrative embodiments of efficiency gain by tuning ofthe antenna structure based on measurements of a different one of theprobes of the antenna structure of FIG. 45;

FIG. 49 depicts an illustrative embodiment of a near field sensor;

FIG. 50 depicts an illustrative embodiment of the near field sensor ofFIG. 49 on a printed circuit board;

FIG. 51 depicts an illustrative embodiment of a first method that can beapplied to the subject disclosure;

FIG. 52 depicts an illustrative embodiment of a near field sensor;

FIG. 53 depicts an illustrative embodiment of a phase detector;

FIG. 54 depict illustrative phase error versus frequency and power levelplots resulting from the near field sensor embodiment of FIG. 52;

FIG. 55 depicts an illustrative embodiment of a second method that canbe applied to the subject disclosure;

FIG. 56 depicts an illustrative embodiment of a reactance sensor;

FIG. 57 depicts an illustrative embodiment of a third method that can beapplied to the subject disclosure;

FIG. 58 depicts an illustrative embodiment of a communication device;and

FIG. 59 is a diagrammatic representation of a machine in the form of acomputer system within which a set of instructions, when executed, maycause the machine to perform any one or more of the methods describedherein.

DETAILED DESCRIPTION

The subject disclosure describes, among other things, illustrativeembodiments for monitoring changes in an operating frequency of anantenna and adjusting the operating frequency of the antenna to mitigatesuch changes. Other embodiments are described in the subject disclosure.

One embodiment of the subject disclosure includes a method formeasuring, by a circuit, from a first probe a first magnitude ofradiated energy by an antenna, where the first probe is placed near theantenna, obtaining, by the circuit, a second magnitude of a signalsupplied to the antenna, comparing, by the circuit, the first and thesecond magnitudes, detecting, by the circuit, an offset in an operatingfrequency of the antenna based on a difference between the first and thesecond magnitudes, and adjusting, by the circuit, the operatingfrequency of the antenna to mitigate the offset in the operatingfrequency of the antenna.

One embodiment of the subject disclosure includes an antenna structurehaving a first antenna element for receiving and transmitting radiofrequency signals within an operating frequency range, a first aperturetuner for adjusting an operating frequency of the antenna element, and afirst near field sensor for sensing radiated energy from the firstantenna element. The first near field sensor, the first antenna element,and the first aperture tuner can be coupled to a circuit that performsoperations comprising measuring from the first near field sensor a firstmagnitude of radiated energy by the first antenna element, obtaining asecond magnitude of a signal supplied to the first antenna element,comparing the first and the second magnitudes, detecting a change in anoperating frequency of the first antenna element based on a differencebetween the first and the second magnitudes, and directing the firstaperture tuner to adjust the operating frequency of the first antennaelement to counter the change in the operating frequency of the firstantenna element.

One embodiment of the subject disclosure includes a communication devicehaving a near field sensor coupled to the antenna structure, and acircuit coupled to the antenna structure and the near field sensor. Thecircuit can perform operations including measuring from the near fieldsensor a first magnitude of radiated energy by the antenna structure,obtaining a second magnitude of a signal supplied to the antennastructure by a transmitter circuit, comparing the first and the secondmagnitudes, detecting an offset in an operating frequency of the antennastructure based on a difference between the first and the secondmagnitudes, and adjusting the operating frequency of the antennastructure to mitigate the offset.

One embodiment of the subject disclosure includes a method formeasuring, by a circuit, from a first probe a first phase of radiatedenergy by an antenna, wherein the first probe is placed near theantenna, measuring, by the circuit, from a second probe a second phaseof a transmit signal supplied to the antenna, wherein the second probeis placed in a transmission path of the transmit signal, comparing, bythe circuit, the first and the second phases to generate a phasedifferential, detecting, by the circuit, an offset in an operatingfrequency of the antenna based on the phase differential, and adjusting,by the circuit, the operating frequency of the antenna to mitigate theoffset in the operating frequency of the antenna.

One embodiment of the subject disclosure includes an antenna structureincluding a first antenna element, a first aperture tuner for adjustingan operating frequency of the antenna element, a probe, and a first nearfield sensor for sensing radiated energy from the first antenna element.The first near field sensor and the first aperture tuner can be coupledto a circuit that performs operations including measuring from the firstnear field sensor a first phase of radiated energy by the first antennaelement, measuring from the probe a second phase of a first signalsupplied to the first antenna element, comparing the first and thesecond phases to generate a first phase differential, detecting a changein an operating frequency of the first antenna element based on thephase differential, and directing the first aperture tuner to adjust theoperating frequency of the first antenna element according to the firstphase differential.

One embodiment of the subject disclosure includes a communication devicehaving an antenna structure, a near field sensor, a probe, and a circuitcoupled to the antenna structure and the near field sensor. The circuitcan perform operations including measuring from the near field sensor afirst signal representing radiated energy from the antenna structure,measuring from the probe a second signal supplied to the antennastructure, determining a phase differential from a first phase of thefirst signal and a second phase of the second signal, detecting afrequency offset of the antenna structure based on the phasedifferential, and adjusting an operating frequency of the antennastructure to mitigate the frequency offset.

One embodiment of the subject disclosure includes a method formeasuring, by a circuit, a change in reactance of an antenna,determining, by the circuit, a frequency offset of the antenna based ona change in an operating frequency of the antenna according to thechange in reactance of the antenna, and adjusting, by the circuit, theoperating frequency of the antenna to mitigate the frequency offset ofthe antenna.

One embodiment of the subject disclosure includes an antenna structurehaving a first antenna element, and a sensor coupled to the firstantenna element. The sensor can be coupled to a circuit that performsoperations including measuring from the sensor a change in a reactanceof the antenna, obtaining impedance characteristics of the antenna,determining a change in an operating frequency of the antenna accordingto the change in reactance of the antenna and the impedancecharacteristics of the antenna, and adjusting the operating frequency ofthe antenna to counteract the change in the operating frequency of theantenna.

One embodiment of the subject disclosure includes a communication devicehaving an antenna structure, a sensor, and a circuit coupled to thesensor. The circuit can perform operations including measuring a changein a reactance of the antenna, determining a frequency offset of theantenna according to the change in reactance of the antenna, andadjusting an operating frequency of the antenna to reduce the frequencyoffset of the antenna.

Antenna structures in accordance with various embodiments of thedisclosure are particularly useful in communications devices thatrequire multiple antennas to be packaged close together (e.g., less thana quarter of a wavelength apart), including in devices where more thanone antenna is used simultaneously and within the same frequency band ormultiple frequency bands in cases where carrier aggregation is required.Common examples of such devices in which the antenna structures can beused include portable communications products such as cellular handsets,PDAs, smart phones, tablets, and wireless networking devices or datacards for PCs or other equipment integrated communication devices suchas automobiles, trucks, or other vehicle categories. The antennastructures are also useful with system architectures such as MIMO andstandard protocols for mobile wireless communications devices (such as802.11n for wireless LAN, and 3G and 4G data communications such as802.16e (WiMAX), HSDPA, 1xEVDO, LTE) that require multiple antennasoperating simultaneously. The embodiments of the subject disclosure canbe applied to future generations of wireless communication protocolssuch as 5G.

FIGS. 1A-1G illustrate the operation of an antenna structure 100. FIG.1A schematically illustrates the antenna structure 100 having twoparallel antennas, in particular parallel dipoles 102, 104, of length L.The dipoles 102, 104 are separated by a distance d, and are notconnected by any connecting element. The dipoles 102, 104 have afundamental resonant frequency that corresponds approximately to L=λ/2.Each dipole is connected to an independent transmit/receive system,which can operate at the same frequency. This system connection can havethe same characteristic impedance z₀ for both antennas, which in thisexample is 50 ohms.

When one dipole is transmitting a signal, some of the signal beingtransmitted by the dipole will be coupled directly into the neighboringdipole. The maximum amount of coupling generally occurs near thehalf-wave resonant frequency of the individual dipole and generallyincreases as the separation distance d is made smaller. For example, ford<λ/3, the magnitude of coupling is greater than 0.1 or −10 dB, and ford<λ/8, the magnitude of the coupling is greater than −5 dB.

It is desirable to have no coupling (i.e., complete isolation) or toreduce the coupling (i.e., at least reduced isolation) between theantennas. If the coupling is, e.g., −10 dB, 10 percent of the transmitpower is lost due to that amount of power being directly coupled intothe neighboring antenna. There may also be detrimental system effectssuch as saturation or desensitization of a receiver connected to theneighboring antenna or degradation of the performance of a transmitterconnected to the neighboring antenna. Currents induced on theneighboring antenna distort the gain pattern compared to that generatedby an individual dipole. This effect is known to reduce the correlationbetween the gain patterns produced by the dipoles. Thus, while couplingmay provide some pattern diversity, it has detrimental system impacts asdescribed above.

Because of the close coupling, the antennas do not act independently andcan be considered an antenna system having two pairs of terminals orports that correspond to two different gain patterns. Use of either portinvolves substantially the entire structure including both dipoles. Theparasitic excitation of the neighboring dipole enables diversity to beachieved at close dipole spacing, but currents excited on the dipolepass through the source impedance, and therefore manifest mutualcoupling between ports.

FIG. 1C illustrates a model dipole pair corresponding to the antennastructure 100 shown in FIG. 1 used for simulations. In this example, thedipoles 102, 104 have a square cross section of 1 mm×1 mm and length (L)of 56 mm. These dimensions yield a center resonant frequency of 2.45 GHzwhen attached to a 50-ohm source. The free-space wavelength at thisfrequency is 122 mm. A plot of the scattering parameters S11 and S21 fora separation distance (d) of 10 mm, or approximately λ/12, is shown inFIG. 1D. Due to symmetry and reciprocity, S22=S11 and S12=S21. Forsimplicity, only S11 and S21 are shown and discussed. In thisconfiguration, the coupling between dipoles as represented by S21reaches a maximum of −3.7 dB.

FIG. 1E shows the ratio (identified as “Magnitude I2/I1” in the figure)of the vertical current on dipole 104 of the antenna structure to thaton dipole 102 under the condition in which port 106 is excited and port108 is passively terminated. The frequency at which the ratio ofcurrents (dipole 104/dipole 102) is a maximum corresponds to thefrequency of 180 degree phase differential between the dipole currentsand is just slightly higher in frequency than the point of maximumcoupling shown in FIG. 1D.

FIG. 1F shows azimuthal gain patterns for several frequencies withexcitation of port 106. The patterns are not uniformly omni-directionaland change with frequency due to the changing magnitude and phase of thecoupling. Due to symmetry, the patterns resulting from excitation ofport 108 would be the mirror image of those for port 106. Therefore, themore asymmetrical the pattern is from left to right, the more diversethe patterns are in terms of gain magnitude.

Calculation of the correlation coefficient between patterns provides aquantitative characterization of the pattern diversity. FIG. 1G showsthe calculated correlation between port 106 and port 108 antennapatterns. The correlation is much lower than is predicted by Clark'smodel for ideal dipoles. This is due to the differences in the patternsintroduced by the mutual coupling.

FIGS. 2A-2F illustrate the operation of an exemplary two port antennastructure 200 in accordance with one or more embodiments of thedisclosure. The two port antenna structure 200 includes twoclosely-spaced resonant antenna elements 202, 204 and provides both lowpattern correlation and low coupling between ports 206, 208. FIG. 2Aschematically illustrates the two port antenna structure 200. Thisstructure is similar to the antenna structure 100 comprising the pair ofdipoles shown in FIG. 1B, but additionally includes horizontalconductive connecting elements 210, 212 between the dipoles on eitherside of the ports 206, 208. The two ports 206, 208 are located in thesame locations as with the FIG. 1 antenna structure. When one port isexcited, the combined structure exhibits a resonance similar to that ofthe unattached pair of dipoles, but with a significant reduction incoupling and an increase in pattern diversity.

An exemplary model of the antenna structure 200 with a 10 mm dipoleseparation is shown in FIG. 2B. This structure has generally the samegeometry as the antenna structure 100 shown in FIG. 1C, but with theaddition of the two horizontal connecting elements 210, 212 electricallyconnecting the antenna elements slightly above and below the ports. Thisstructure shows a strong resonance at the same frequency as unattacheddipoles, but with very different scattering parameters as shown in FIG.2C. There is a deep drop-out in coupling, below −20 dB, and a shift inthe input impedance as indicated by S11. In this example, the bestimpedance match (S11 minimum) does not coincide with the lowest coupling(S21 minimum). A matching network can be used to improve the inputimpedance match and still achieve very low coupling as shown in FIG. 2D.In this example, a lumped element matching network comprising a seriesinductor followed by a shunt capacitor was added between each port andthe structure.

FIG. 2E shows the ratio (indicated as “Magnitude I2/I1” in the figure)of the current on dipole element 204 to that on dipole element 202resulting from excitation of port 206. This plot shows that below theresonant frequency, the currents are actually greater on dipole element204. Near resonance, the currents on dipole element 204 begin todecrease relative to those on dipole element 202 with increasingfrequency. A point of low coupling (2.44 GHz in this case) occurs nearthe frequency where currents on both dipole elements are generally equalin magnitude. At this frequency, the phase of the currents on dipoleelement 204 lag those of dipole element 202 by approximately 160degrees.

Unlike the FIG. 1C dipoles without connecting elements, the currents onantenna element 204 of the FIG. 2B combined antenna structure 200 arenot forced to pass through the terminal impedance of port 208. Instead aresonant mode is produced where the current flows down antenna element204, across the connecting element 210, 212, and up antenna element 202as indicated by the arrows shown on FIG. 2A. (Note that this currentflow is representative of one half of the resonant cycle; during theother half, the current directions are reversed). The resonant mode ofthe combined structure features the following: (1) the currents onantenna element 204 largely bypass port 208, thereby allowing for highisolation between the ports 206, 208, and (2) the magnitude of thecurrents on both antenna elements 202,204 are approximately equal, whichallows for dissimilar and uncorrelated gain patterns as described infurther detail below.

Because the magnitude of currents is nearly equal on the antennaelements, a much more directional pattern is produced (as shown on FIG.2F) than in the case of the FIG. 1C antenna structure 100 withunattached dipoles. When the currents are equal, the condition fornulling the pattern in the x (or phi=0) direction is for the phase ofcurrents on dipole 204 to lag those of dipole 202 by the quantity π-kd(where k=2π/λ, and λ is the effective wavelength). Under this condition,fields propagating in the phi=0 direction from dipole 204 will be 180degrees out of phase with those of dipole 202, and the combination ofthe two will therefore have a null in the phi=0 direction.

In the model example of FIG. 2B, d is 10 mm or an effective electricallength of λ/12. In this case, kd equates π/6 or 30 degrees, and so thecondition for a directional azimuthal radiation pattern with a nulltowards phi=0 and maximum gain towards phi=180 is for the current ondipole 204 to lag those on dipole 202 by 150 degrees. At resonance, thecurrents pass close to this condition (as shown in FIG. 2E), whichexplains the directionality of the patterns. In the case of theexcitation of port 204, the radiation patterns are the mirror oppositeof those of FIG. 2F, and maximum gain is in the phi=0 direction. Thedifference in antenna patterns produced from the two ports has anassociated low predicted envelope correlation as shown on FIG. 2G. Thusthe combined antenna structure has two ports that are isolated from eachother and produce gain patterns of low correlation.

Accordingly, the frequency response of the coupling is dependent on thecharacteristics of the connecting elements 210, 212, including theirimpedance and electrical length. In accordance with one or moreembodiments of the disclosure, the frequency or bandwidth over which adesired amount of isolation can be maintained is controlled byappropriately configuring the connecting elements. One way to configurethe cross connection is to change the physical length of the connectingelement. An example of this is shown by the multimode antenna structure300 of FIG. 3A where a meander has been added to the cross connectionpath of the connecting elements 310, 312. This has the general effect ofincreasing both the electrical length and the impedance of theconnection between the two antenna elements 302, 304. Performancecharacteristics of this structure including scattering parameters,current ratios, gain patterns, and pattern correlation are shown onFIGS. 3B, 3C, 3D, and 3E, respectively. In this embodiment, the changein physical length has not significantly altered the resonant frequencyof the structure, but there is a significant change in S21, with largerbandwidth and a greater minimum value than in structures without themeander. Thus, it is possible to optimize or improve the isolationperformance by altering the electrical characteristic of the connectingelements.

Exemplary multimode antenna structures in accordance with variousembodiments of the disclosure can be designed to be excited from aground or counterpoise 402 (as shown by antenna structure 400 in FIG.4), or as a balanced structure (as shown by antenna structure 500 inFIG. 5). In either case, each antenna structure includes two or moreantenna elements (402, 404 in FIGS. 4, and 502, 504 in FIG. 5) and oneor more electrically conductive connecting elements (406 in FIGS. 4, and506, 508 in FIG. 5). For ease of illustration, only a two-port structureis illustrated in the example diagrams. However, it is possible toextend the structure to include more than two ports in accordance withvarious embodiments of the disclosure. A signal connection to theantenna structure, or port (418, 412 in FIGS. 4 and 510, 512 in FIG. 5),is provided at each antenna element. The connecting element provideselectrical connection between the two antenna elements at the frequencyor frequency range of interest. Although the antenna is physically andelectrically one structure, its operation can be explained byconsidering it as two independent antennas. For antenna structures notincluding a connecting element such as antenna structure 100, port 106of that structure can be said to be connected to antenna 102, and port108 can be said to be connected to antenna 104. However, in the case ofthis combined structure such as antenna structure 400, port 418 can bereferred to as being associated with one antenna mode, and port 412 canbe referred to as being associated with another antenna mode.

The antenna elements are designed to be resonant at the desiredfrequency or frequency range of operation. The lowest order resonanceoccurs when an antenna element has an electrical length of one quarterof a wavelength. Thus, a simple element design is a quarter-wavemonopole in the case of an unbalanced configuration. It is also possibleto use higher order modes. For example, a structure formed fromquarter-wave monopoles also exhibits dual mode antenna performance withhigh isolation at a frequency of three times the fundamental frequency.Thus, higher order modes may be exploited to create a multiband antenna.Similarly, in a balanced configuration, the antenna elements can becomplementary quarter-wave elements as in a half-wave center-fed dipole.However, the antenna structure can also be formed from other types ofantenna elements that are resonant at the desired frequency or frequencyrange. Other possible antenna element configurations include, but arenot limited to, helical coils, wideband planar shapes, chip antennas,meandered shapes, loops, and inductively shunted forms such as PlanarInverted-F Antennas (PIFAs).

The antenna elements of an antenna structure in accordance with one ormore embodiments of the disclosure need not have the same geometry or bethe same type of antenna element. The antenna elements should each haveresonance at the desired frequency or frequency range of operation.

In accordance with one or more embodiments of the disclosure, theantenna elements of an antenna structure have the same geometry. This isgenerally desirable for design simplicity, especially when the antennaperformance requirements are the same for connection to either port.

The bandwidth and resonant frequencies of the combined antenna structurecan be controlled by the bandwidth and resonance frequencies of theantenna elements. Thus, broader bandwidth elements can be used toproduce a broader bandwidth for the modes of the combined structure asillustrated, e.g., in FIGS. 6A, 6B, and 6C. FIG. 6A illustrates amultimode antenna structure 600 including two dipoles 602, 604 connectedby connecting elements 606, 608. The dipoles 602, 604 each have a width(W) and a length (L) and are spaced apart by a distance (d). FIG. 6Billustrates the scattering parameters for the structure having exemplarydimensions: W=1 mm, L=57.2 mm, and d=10 mm. FIG. 6C illustrates thescattering parameters for the structure having exemplary dimensions:W=10 mm, L=50.4 mm, and d=10 mm. As shown, increasing W from 1 mm to 10mm, while keeping the other dimensions generally the same, results in abroader isolation bandwidth and impedance bandwidth for the antennastructure.

It has also been found that increasing the separation between theantenna elements increases the isolation bandwidth and the impedancebandwidth for an antenna structure.

In general, the connecting element is in the high-current region of thecombined resonant structure. It may therefore be desirable for theconnecting element to have a high conductivity.

The ports are located at the feed points of the antenna elements as theywould be if they were operated as separate antennas. Matching elementsor structures may be used to match the port impedance to the desiredsystem impedance.

In accordance with one or more embodiments of the disclosure, themultimode antenna structure can be a planar structure incorporated,e.g., into a printed circuit board, as shown as FIG. 7. In this example,the antenna structure 700 includes antenna elements 702, 704 connectedby a connecting element 706 at ports 708, 710. The antenna structure isfabricated on a printed circuit board substrate 712. The antennaelements shown in the figure are simple quarter-wave monopoles. However,the antenna elements can be any geometry that yields an equivalenteffective electrical length.

In accordance with one or more embodiments of the disclosure, antennaelements with dual resonant frequencies can be used to produce acombined antenna structure with dual resonant frequencies and hence dualoperating frequencies. FIG. 8A shows an exemplary model of a multimodedipole structure 800 where the dipole antenna elements 802, 804 aresplit into two fingers 806, 808 and 810, 812, respectively, of unequallength. The dipole antenna elements have resonant frequencies associatedwith each the two different finger lengths and accordingly exhibit adual resonance. Similarly, the multimode antenna structure usingdual-resonant dipole arms exhibits two frequency bands where highisolation (or small S21) is obtained as shown in FIG. 8B.

In accordance with one or more embodiments of the disclosure, amultimode antenna structure 900 shown in FIG. 9 is provided havingvariable length antenna elements 902, 904 forming a tunable antenna.This may be done by changing the effective electrical length of theantenna elements by a controllable device such as an RF switch 906, 908at each antenna element 902, 904. In this example, the switch may beopened (by operating the controllable device) to create a shorterelectrical length (for higher frequency operation) or closed to create alonger electrical length (for lower frequency of operation). Theoperating frequency band for the antenna structure 900, including thefeature of high isolation, can be tuned by tuning both antenna elementsin concert. This approach may be used with a variety of methods ofchanging the effective electrical length of the antenna elementsincluding, e.g., using a controllable dielectric material, loading theantenna elements with a variable capacitor such as amicroelectromechanical systems (MEMs) device, varactor, or tunabledielectric capacitor, and switching on or off parasitic elements.

In accordance with one or more embodiments of the disclosure, theconnecting element or elements provide an electrical connection betweenthe antenna elements with an electrical length approximately equal tothe electrical distance between the elements. Under this condition, andwhen the connecting elements are attached at the port ends of theantenna elements, the ports are isolated at a frequency near theresonance frequency of the antenna elements. This arrangement canproduce nearly perfect isolation at particular frequency.

Alternately, as previously discussed, the electrical length of theconnecting element may be increased to expand the bandwidth over whichisolation exceeds a particular value. For example, a straight connectionbetween antenna elements may produce a minimum S21 of −25 dB at aparticular frequency and the bandwidth for which S21<−10 dB may be 100MHz. By increasing the electrical length, a new response can be obtainedwhere the minimum S21 is increased to −15 dB but the bandwidth for whichS21<−10 dB may be increased to 150 MHz.

Various other multimode antenna structures in accordance with one ormore embodiments of the disclosure are possible. For example, theconnecting element can have a varied geometry or can be constructed toinclude components to vary the properties of the antenna structure.These components can include, e.g., passive inductor and capacitorelements, resonator or filter structures, or active components such asphase shifters.

In accordance with one or more embodiments of the disclosure, theposition of the connecting element along the length of the antennaelements can be varied to adjust the properties of the antennastructure. The frequency band over which the ports are isolated can beshifted upward in frequency by moving the point of attachment of theconnecting element on the antenna elements away from the ports andtowards the distal end of the antenna elements. FIGS. 10A and 10Billustrate multimode antenna structures 1000, 1002, respectively, eachhaving a connecting element electrically connected to the antennaelements. In the FIG. 10A antenna structure 1000, the connecting element1004 is located in the structure such that the gap between theconnecting element 1004 and the top edge of the ground plane 1006 is 3mm. FIG. 10C shows the scattering parameters for the structure showingthat high isolation is obtained at a frequency of 1.15 GHz in thisconfiguration. A shunt capacitor/series inductor matching network isused to provide the impedance match at 1.15 GHz. FIG. 10D shows thescattering parameters for the structure 1002 of FIG. 10B, where the gapbetween the connecting element 1008 and the top edge 1010 of the groundplane is 19 mm. The antenna structure 1002 of FIG. 10B exhibits anoperating band with high isolation at approximately 1.50 GHz.

FIG. 11 schematically illustrates a multimode antenna structure 1100 inaccordance with one or more further embodiments of the disclosure. Theantenna structure 1100 includes two or more connecting elements 1102,1104, each of which electrically connects the antenna elements 1106,1108. (For ease of illustration, only two connecting elements are shownin the figure. It should be understood that use of more than twoconnecting elements is also contemplated.) The connecting elements 1102,1104 are spaced apart from each other along the antenna elements 1106,1108. Each of the connecting elements 1102, 1104 includes a switch 1112,1110. Peak isolation frequencies can be selected by controlling theswitches 1110, 1112. For example, a frequency f1 can be selected byclosing switch 1110 and opening switch 1112. A different frequency f2can be selected by closing switch 1112 and opening switch 1110.

FIG. 12 illustrates a multimode antenna structure 1200 in accordancewith one or more alternate embodiments of the disclosure. The antennastructure 1200 includes a connecting element 1202 having a filter 1204operatively coupled thereto. The filter 1204 can be a low pass or bandpass filter selected such that the connecting element connection betweenthe antenna elements 1206, 1208 is only effective within the desiredfrequency band, such as the high isolation resonance frequency. Athigher frequencies, the structure will function as two separate antennaelements that are not coupled by the electrically conductive connectingelement, which is open circuited.

FIG. 13 illustrates a multimode antenna structure 1300 in accordancewith one or more alternate embodiments of the disclosure. The antennastructure 1300 includes two or more connecting elements 1302, 1304,which include filters 1306, 1308, respectively. (For ease ofillustration, only two connecting elements are shown in the figure. Itshould be understood that use of more than two connecting elements isalso contemplated.) In one possible embodiment, the antenna structure1300 has a low pass filter 1308 on the connecting element 1304 (which iscloser to the antenna ports) and a high pass filter 1306 on theconnecting element 1302 in order to create an antenna structure with twofrequency bands of high isolation, i.e., a dual band structure.

FIG. 14 illustrates a multimode antenna structure 1400 in accordancewith one or more alternate embodiments of the disclosure. The antennastructure 1400 includes one or more connecting elements 1402 having atunable element 1406 operatively connected thereto. The antennastructure 1400 also includes antenna elements 1408, 1410. The tunableelement 1406 alters the delay or phase of the electrical connection orchanges the reactive impedance of the electrical connection. Themagnitude of the scattering parameters S21/S12 and a frequency responseare affected by the change in electrical delay or impedance and so anantenna structure can be adapted or generally optimized for isolation atspecific frequencies using the tunable element 1406.

FIG. 15 illustrates a multimode antenna structure 1500 in accordancewith one or more alternate embodiments of the disclosure. The multimodeantenna structure 1500 can be used, e.g., in a WIMAX USB dongle. Theantenna structure 1500 can be configured for operation, e.g., in WiMAXbands from 2300 to 2700 MHz.

The antenna structure 1500 includes two antenna elements 1502, 1504connected by a conductive connecting element 1506. The antenna elementsinclude slots to increase the electrical length of the elements toobtain the desired operating frequency range. In this example, theantenna structure is optimized for a center frequency of 2350 MHz. Thelength of the slots can be reduced to obtain higher center frequencies.The antenna structure is mounted on a printed circuit board assembly1508. A two-component lumped element match is provided at each antennafeed.

The antenna structure 1500 can be manufactured, e.g., by metal stamping.It can be made, e.g., from 0.2 mm thick copper alloy sheet. The antennastructure 1500 includes a pickup feature 1510 on the connecting elementat the center of mass of the structure, which can be used in anautomated pick-and-place assembly process. The antenna structure is alsocompatible with surface-mount reflow assembly.

FIG. 16 illustrates a multimode antenna structure 1600 in accordancewith one or more alternate embodiments of the disclosure. As withantenna structure 1500 of FIG. 15, the antenna structure 1600 can alsobe used, e.g., in a WIMAX USB dongle. The antenna structure can beconfigured for operation, e.g., in WiMAX bands from 2300 to 2700 MHz.

The antenna structure 1600 includes two antenna elements 1602, 1604,each comprising a meandered monopole. The length of the meanderdetermines the center frequency. The exemplary design shown in thefigure is optimized for a center frequency of 2350 MHz. To obtain highercenter frequencies, the length of the meander can be reduced.

A connecting element 1606 electrically connects the antenna elements. Atwo-component lumped element match is provided at each antenna feed.

The antenna structure can be fabricated, e.g., from copper as a flexibleprinted circuit (FPC) mounted on a plastic carrier 1608. The antennastructure can be created by the metalized portions of the FPC. Theplastic carrier provides mechanical support and facilitates mounting toa PCB assembly 1610. Alternatively, the antenna structure can be formedfrom sheet-metal.

FIG. 17 illustrates a multimode antenna structure 1700 in accordancewith another embodiment of the disclosure. This antenna design can beused, e.g., for USB, Express 34, and Express 54 data card formats. Theexemplary antenna structure shown in the figure is designed to operateat frequencies from 2.3 to 6 GHz. The antenna structure can befabricated, e.g., from sheet-metal or by FPC over a plastic carrier1702.

FIG. 18A illustrates a multimode antenna structure 1800 in accordancewith another embodiment of the disclosure. The antenna structure 1800comprises a three mode antenna with three ports. In this structure,three monopole antenna elements 1802, 1804, 1806 are connected using aconnecting element 1808 comprising a conductive ring that connectsneighboring antenna elements. The antenna elements are balanced by acommon counterpoise, or sleeve 1810, which is a single hollow conductivecylinder. The antenna has three coaxial cables 1812, 1814, 1816 forconnection of the antenna structure to a communications device. Thecoaxial cables 1812, 1814, 1816 pass through the hollow interior of thesleeve 1810. The antenna assembly may be constructed from a singleflexible printed circuit wrapped into a cylinder and may be packaged ina cylindrical plastic enclosure to provide a single antenna assemblythat takes the place of three separate antennas. In one exemplaryarrangement, the diameter of the cylinder is 10 mm and the overalllength of the antenna is 56 mm so as to operate with high isolationbetween ports at 2.45 GHz. This antenna structure can be used, e.g.,with multiple antenna radio systems such as MIMO or 802.11N systemsoperating in the 2.4 to 2.5 GHz bands. In addition to port to portisolation, each port advantageously produces a different gain pattern asshown on FIG. 18B. While this is one specific example, it is understoodthat this structure can be scaled to operate at any desired frequency.It is also understood that methods for tuning, manipulating bandwidth,and creating multiband structures described previously in the context oftwo-port antennas can also apply to this multiport structure.

While the above embodiment is shown as a true cylinder, it is possibleto use other arrangements of three antenna elements and connectingelements that produce the same advantages. This includes, but is notlimited to, arrangements with straight connections such that theconnecting elements form a triangle, or another polygonal geometry. Itis also possible to construct a similar structure by similarlyconnecting three separate dipole elements instead of three monopoleelements with a common counterpoise. Also, while symmetric arrangementof antenna elements advantageously produces equivalent performance fromeach port, e.g., same bandwidth, isolation, impedance matching, it isalso possible to arrange the antenna elements asymmetrically or withunequal spacing depending on the application.

FIG. 19 illustrates use of a multimode antenna structure 1900 in acombiner application in accordance with one or more embodiments of thedisclosure. As shown in the figure, transmit signals may be applied toboth antenna ports of the antenna structure 1900 simultaneously. In thisconfiguration, the multimode antenna can serve as both antenna and poweramplifier combiner. The high isolation between antenna ports restrictsinteraction between the two amplifiers 1902, 1904, which is known tohave undesirable effects such as signal distortion and loss ofefficiency. Optional impedance matching at 1906 can be provided at theantenna ports.

FIGS. 20A and 20B illustrate a multimode antenna structure 2000 inaccordance with one or more alternate embodiments of the subjectdisclosure. The antenna structure 2000 can also be used, e.g., in aWiMAX USB or ExpressCard/34 device. The antenna structure can beconfigured for operation, e.g., in WiMAX bands from 2300 to 6000 MHz.

The antenna structure 2000 includes two antenna elements 2001, 2004,each comprising a broad monopole. A connecting element 2002 electricallyconnects the antenna elements. Slots (or other cut-outs) 2005 are usedto improve the input impedance match above 5000 MHz. The exemplarydesign shown in the figure is optimized to cover frequencies from 2300to 6000 MHz.

The antenna structure 2000 can be manufactured, e.g., by metal stamping.It can be made, e.g., from 0.2 mm thick copper alloy sheet. The antennastructure 2000 includes a pickup feature 2003 on the connecting element2002 generally at the center of mass of the structure, which can be usedin an automated pick-and-place assembly process. The antenna structureis also compatible with surface-mount reflow assembly. Feed points 2006of the antenna provide the points of connection to the radio circuitryon a PCB, and also serve as a support for structural mounting of theantenna to the PCB. Additional contact points 2007 provide structuralsupport.

FIG. 20C illustrates a test assembly 2010 used to measure theperformance of antenna 2000. The figure also shows the coordinatereference for far-field patterns. Antenna 2000 is mounted on a 30×88 mmPCB 2011 representing an ExpressCard/34 device. The grounded portion ofthe PCB 2011 is attached to a larger metal sheet 2012 (having dimensionsof 165×254 mm in this example) to represent a counterpoise size typicalof a notebook computer. Test ports 2014, 2016 on the PCB 2011 areconnected to the antenna through 50-ohm striplines.

FIG. 20D shows the VSWR measured at test ports 2014, 2016. FIG. 20Eshows the coupling (S21 or S12) measured between the test ports. TheVSWR and coupling are advantageously low across the broad range offrequencies, e.g., 2300 to 6000 MHz. FIG. 20F shows the measuredradiation efficiency referenced from the test ports 2014 (Port 1), 2016(Port 2). FIG. 20G shows the calculated correlation between theradiation patterns produced by excitation of test port 2014 (Port 1)versus those produced by excitation of test port 2016 (Port 2). Theradiation efficiency is advantageously high while the correlationbetween patterns is advantageously low at the frequencies of interest.FIG. 20H shows far field gain patterns by excitation of test port 2014(Port 1) or test port 2016 (Port 2) at a frequency of 2500 MHz. FIGS.20I and 20J show the same pattern measurements at frequencies of 3500and 5200 MHz, respectively. The patterns resulting from test port 2014(Port 1) are different and complementary to those of test port 2016(Port 2) in the φ=0 or XZ plane and in the θ=90 or XY plane.

FIGS. 21A and 21B illustrate a multimode antenna structure 2100 inaccordance with one or more alternate embodiments of the subjectdisclosure. The antenna structure 2100 can also be used, e.g., in aWiMAX USB dongle. The antenna structure can be configured for operation,e.g., in WiMAX bands from 2300 to 2400 MHz.

The antenna structure 2100 includes two antenna elements 2102, 2104,each comprising a meandered monopole. The length of the meanderdetermines the center frequency. Other tortuous configurations such as,e.g., helical coils and loops, can also be used to provide a desiredelectrical length. The exemplary design shown in the figure is optimizedfor a center frequency of 2350 MHz. A connecting element 2106 (shown inFIG. 21B) electrically connects the antenna elements 2102, 2104. Atwo-component lumped element match is provided at each antenna feed.

The antenna structure can be fabricated, e.g., from copper as a flexibleprinted circuit (FPC) 2103 mounted on a plastic carrier 2101. Theantenna structure can be created by the metalized portions of the FPC2103. The plastic carrier 2101 provides mounting pins or pips 2107 forattaching the antenna to a PCB assembly (not shown) and pips 2105 forsecuring the FPC 2103 to the carrier 2101. The metalized portion of 2103includes exposed portions or pads 2108 for electrically contacting theantenna to the circuitry on the PCB.

To obtain higher center frequencies, the electrical length of theelements 2102, 2104 can be reduced. FIGS. 22A and 22B illustrate amultimode antenna structure 2200, the design of which is optimized for acenter frequency of 2600 MHz. The electrical length of the elements2202, 2204 is shorter than that of elements 2102, 2104 of FIGS. 21A and21B because metallization at the end of the elements 2202, 2204 has beenremoved, and the width of the of the elements at feed end has beenincreased.

FIG. 23A illustrates a test assembly 2300 using antenna 2100 of FIGS.21A and 21B along with the coordinate reference for far-field patterns.FIG. 23B shows the VSWR measured at test ports 2302 (Port 1), 2304 (Port2). FIG. 23C shows the coupling (S21 or S12) measured between the testports 2302 (Port 1), 2304 (Port 2). The VSWR and coupling areadvantageously low at the frequencies of interest, e.g., 2300 to 2400MHz. FIG. 23D shows the measured radiation efficiency referenced fromthe test ports. FIG. 23E shows the calculated correlation between theradiation patterns produced by excitation of test port 2302 (Port 1)versus those produced by excitation of test port 2304 (Port 2). Theradiation efficiency is advantageously high while the correlationbetween patterns is advantageously low at the frequencies of interest.FIG. 23F shows far field gain patterns by excitation of test port 2302(Port 1) or test port 2304 (Port 2) at a frequency of 2400 MHz. Thepatterns resulting from test port 2302 (Port 1) are different andcomplementary to those of test port 2304 (Port 2) in the 4=0 or XZ planeand in the θ=90 or XY plane.

FIG. 23G shows the VSWR measured at the test ports of assembly 2300 withantenna 2200 in place of antenna 2100. FIG. 23H shows the coupling (S21or S12) measured between the test ports. The VSWR and coupling areadvantageously low at the frequencies of interest, e.g. 2500 to 2700MHz. FIG. 23I shows the measured radiation efficiency referenced fromthe test ports. FIG. 23J shows the calculated correlation between theradiation patterns produced by excitation of test port 2302 (Port 1)versus those produced by excitation of test port 2304 (Port 2). Theradiation efficiency is advantageously high while the correlationbetween patterns is advantageously low at the frequencies of interest.FIG. 23K shows far field gain patterns by excitation of test port 2302(Port 1) or test port 2304 (Port 2) at a frequency of 2600 MHz. Thepatterns resulting from test port 2302 (Port 1) are different andcomplementary to those of test port 2304 (Port 2) in the 4=0 or XZ planeand in the θ=90 or XY plane.

One or more further embodiments of the subject disclosure are directedto techniques for beam pattern control for the purpose of null steeringor beam pointing. When such techniques are applied to a conventionalarray antenna (comprising separate antenna elements that are spaced atsome fraction of a wavelength), each element of the array antenna is fedwith a signal that is a phase shifted version of a reference signal orwaveform. For a uniform linear array with equal excitation, the beampattern produced can be described by the array factor F, which dependson the phase of each individual element and the inter-element elementspacing d.

$F = {A_{0}{\sum\limits_{n = 0}^{N - 1}\; {\exp \left\lbrack {j\; {n\left( {{\beta \; d\mspace{14mu} \cos \mspace{14mu} \theta} + \alpha} \right)}} \right\rbrack}}}$where  β = 2 π/λ, N = Total  #  of  elements, α = phase  shift  between  successive  elements, andθ = angle  from  array  axis

By controlling the phase α to a value α_(i), the maximum value of F canbe adjusted to a different direction θ_(i), thereby controlling thedirection in which a maximum signal is broadcast or received.

The inter-element spacing in conventional array antennas is often on theorder of ¼ wavelength, and the antennas can be closely coupled, havingnearly identical polarization. It is advantageous to reduce the couplingbetween elements, as coupling can lead to several problems in the designand performance of array antennas. For example, problems such as patterndistortion and scan blindness (see Stutzman, Antenna Theory and Design,Wiley 1998, pgs 122-128 and 135-136, and 466-472) can arise fromexcessive inter-element coupling, as well as a reduction of the maximumgain attainable for a given number of elements.

Beam pattern control techniques can be advantageously applied to allmultimode antenna structures described herein having antenna elementsconnected by one or more connecting elements, which exhibit highisolation between multiple feedpoints. The phase between ports at thehigh isolation antenna structure can be used for controlling the antennapattern. It has been found that a higher peak gain is achievable ingiven directions when the antenna is used as a simple beam-forming arrayas a result of the reduced coupling between feedpoints. Accordingly,greater gain can be achieved in selected directions from a highisolation antenna structure in accordance with various embodiments thatutilizes phase control of the carrier signals presented to its feedterminals.

In handset applications where the antennas are spaced at much less than¼ wavelength, mutual coupling effects in conventional antennas reducethe radiation efficiency of the array, and therefore reduce the maximumgain achievable.

By controlling the phase of the carrier signal provided to eachfeedpoint of a high isolation antenna in accordance with variousembodiments, the direction of maximum gain produced by the antennapattern can be controlled. A gain advantage of, e.g., 3 dB obtained bybeam steering is advantageous particularly in portable deviceapplications where the beam pattern is fixed and the device orientationis randomly controlled by the user. As shown, e.g., in the schematicblock diagram of FIG. 24, which illustrates a pattern control apparatus2400 in accordance with various embodiments, a relative phase shift α isapplied by a phase shifter 2402 to the RF signals applied to eachantenna feed 2404, 2408. The signals are fed to respective antenna portsof antenna structure 2410.

The phase shifter 2402 can comprise standard phase shift components suchas, e.g., electrically controlled phase shift devices or standard phaseshift networks.

FIGS. 25A-25G provide a comparison of antenna patterns produced by aclosely spaced 2-D conventional array of dipole antennas and a 2-D arrayof high isolation antennas in accordance with various embodiments of thesubject disclosure for different phase differences a between two feedsto the antennas. In FIGS. 25A-25G, curves are shown for the antennapatterns at θ=90 degrees. The solid lines in the figures represents theantenna pattern produced by the isolated feed single element antenna inaccordance with various embodiments, while the dashed lines representthe antenna pattern produced by two separate monopole conventionalantennas separated by a distance equal to the width of the singleelement isolated feed structure. Therefore, the conventional antenna andthe high isolation antenna are of generally equivalent size.

In all cases shown in the figures, the peak gain produced by the highisolation antenna in accordance with various embodiments produces agreater gain margin when compared to the two separate conventionaldipoles, while providing azimuthal control of the beam pattern. Thisbehavior makes it possible to use the high isolation antenna in transmitor receive applications where additional gain is needed or desired in aparticular direction. The direction can be controlled by adjusting therelative phase between the drivepoint signals. This may be particularlyadvantageous for portable devices needing to direct energy toward areceive point such as, e.g., a base station. The combined high isolationantenna offers greater advantage when compared to two singleconventional antenna elements when phased in a similar fashion.

As shown in FIG. 25A, the combined dipole in accordance with variousembodiments shows greater gain in a uniform azimuth pattern (θ=90) forα=0 (zero degrees phase difference).

As shown in FIG. 25B, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a non-symmetricazimuthal pattern (θ=90 plot for α=30 (30 degrees phase differencebetween feedpoints).

As shown in FIG. 25C, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a shifted azimuthalpattern (θ=90 plot for α=60 (60 degrees phase difference betweenfeedpoints).

As shown in FIG. 25D, the combined dipole in accordance with variousembodiments shows even greater peak gain (at φ=0) with a shiftedazimuthal pattern (θ=90 plot for α=90 (90 degrees phase differencebetween feedpoints).

As shown in FIG. 25E, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a shifted azimuthalpattern (θ=90 plot greater backlobe (at φ=180) for α=120 (120 degreesphase difference between feedpoints).

As shown in FIG. 25F, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a shifted azimuthalpattern (θ=90 plot), even greater backlobe (at φ=180) for α=150 (150degrees phase difference between feedpoints).

As shown in FIG. 25G, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0 &180) with a double lobedazimuthal pattern (θ=90 plot) for α=180 (180 degrees phase differencebetween feedpoints).

FIG. 26 illustrates the ideal gain advantage if the combined highisolation antenna in accordance with one or more embodiments over twoseparate dipoles as a function of the phase angle difference between thefeedpoints for a two feedpoint antenna array.

Further embodiments of the subject disclosure are directed to multimodeantenna structures that provide increased high isolation betweenmulti-band antenna ports operating in close proximity to each other at agiven frequency range. In these embodiments, a band-rejection slot isincorporated in one of the antenna elements of the antenna structure toprovide reduced coupling at the frequency to which the slot is tuned.

FIG. 27A schematically illustrates a simple dual-band branch linemonopole antenna 2700. The antenna 2700 includes a band-rejection slot2702, which defines two branch resonators 2704, 2706. The antenna isdriven by signal generator 2708. Depending on the frequency at which theantenna 2700 is driven, various current distributions are realized onthe two branch resonators 2704, 2706.

The physical dimensions of the slot 2702 are defined by the width Ws andthe length Ls as shown in FIG. 27A. When the excitation frequencysatisfies the condition of Ls=lo/4, the slot feature becomes resonant.At this point the current distribution is concentrated around theshorted section of the slot, as shown in FIG. 27B.

The currents flowing through the branch resonators 2704, 2706 areapproximately equal and oppositely directed along the sides of the slot2702. This causes the antenna structure 2700 to behave in a similarmanner to a spurline band stop filter 2720 (shown schematically in FIG.27C), which transforms the antenna input impedance down significantlylower than the nominal source impedance. This large impedance mismatchresults in a very high VSWR, shown in FIGS. 27D and 27E, and as a resultleads to the desired frequency rejection.

This band-rejection slot technique can be applied to an antenna systemwith two (or more) antennas elements operating in close proximity toeach other where one antenna element needs to pass signals of a desiredfrequency and the other does not. In one or more embodiments, one of thetwo antenna elements includes a band-rejection slot, and the other doesnot. FIG. 28 schematically illustrates an antenna structure 2800, whichincludes a first antenna element 2802, a second antenna element 2804,and a connecting element 2806. The antenna structure 2800 includes ports2808 and 2810 at antenna elements 2802 and 2804, respectively. In thisexample, a signal generator drives the antenna structure 2802 at port2808, while a meter is coupled to the port 2810 to measure current atport 2810. It should be understood, however, that either or both portscan be driven by signal generators. The antenna element 2802 includes aband-rejection slot 2812, which defines two branch resonators 2814,2816. In this embodiment, the branch resonators comprise the maintransmit section of the antenna structure, while the antenna element2804 comprises a diversity receive portion of the antenna structure.

Due to the large mismatch at the port of the antenna element 2802 withthe band-reject slot 2812, the mutual coupling between it and thediversity receive antenna element 2804, which is actually matched at theslot resonant frequency will be quite small and will result inrelatively high isolation.

FIG. 29A is a perspective view of a multimode antenna structure 2900comprising a multi-band diversity receive antenna system that utilizesthe band-rejection slot technique in the GPS band in accordance with oneor more further embodiments of the subject disclosure. (The GPS band is1575.42 MHz with 20 MHz bandwidth.) The antenna structure 2900 is formedon a flex film dielectric substrate 2902, which is formed as a layer ona dielectric carrier 2904. The antenna structure 2900 includes a GPSband rejection slot 2906 on the primary transmit antenna element 2908 ofthe antenna structure 2900. The antenna structure 2900 also includes adiversity receive antenna element 2910, and a connecting element 2912connecting the diversity receive antenna element 2910 and the primarytransmit antenna element 2908. A GPS receiver (not shown) is connectedto the diversity receive antenna element 2910. In order to generallyminimize the antenna coupling from the primary transmit antenna element2908 and to generally maximize the diversity antenna radiationefficiency at these frequencies, the primary antenna element 2908includes the band-rejection slot 2906 and is tuned to an electricalquarter wave length near the center of the GPS band. The diversityreceive antenna element 2910 does not contain such a band rejectionslot, but comprises a GPS antenna element that is properly matched tothe main antenna source impedance so that there will be generallymaximum power transfer between it and the GPS receiver. Although bothantenna elements 2908, 2910 co-exist in close proximity, the high VSWRdue to the slot 2906 at the primary transmit antenna element 2908reduces the coupling to the primary antenna element source resistance atthe frequency to which the slot 2906 is tuned, and therefore providesisolation at the GPS frequency between both antenna elements 2908, 2910.The resultant mismatch between the two antenna elements 2908, 2910within the GPS band is large enough to decouple the antenna elements inorder to meet the isolation requirements for the system design as shownin FIGS. 29B and 29C.

In the antenna structures described herein in accordance with variousembodiments of the subject disclosure, the antenna elements and theconnecting elements can form a single integrated radiating structuresuch that a signal fed to either port excites the entire antennastructure to radiate as a whole, rather than separate radiatingstructures. As such, the techniques described herein provide isolationof the antenna ports without the use of decoupling networks at theantenna feed points.

Other embodiments disclosed herein are directed to an antenna thatseparates the fundamental (low band) resonance from the high bandresonance by using two separate structures, which are connected at thefeedpoint—thus accomplishing the goal of achieving a MIMO or Diversityantenna with each feed exhibiting a multiband capability, and wherebyeach feed is optimally isolated from the opposite feed. By way of anon-limiting illustration, in some implementations, high bandfrequencies can range from 1710 to 2170 MHz, and low band frequenciescan range from 698 to 960 MHz.

In one or more embodiments of the antenna structures described in thesubject disclosure, electrical currents flowing through neighboringantenna elements 3002 and 3004 (see FIG. 30) can be configured to besimilar in magnitude, such that an antenna mode excited by one antennaport (e.g., Port 1) is approximately electrically isolated from anantenna mode excited by another antenna port (e.g., Port 2) at a givendesired signal frequency range. In one embodiment, this can beaccomplished by configuring antennas 3002 and 3004 with a connectingelement 3006 to enable common and difference mode currents, which whensummed together result in some or a substantial amount of isolationbetween ports 1 and 2. Configuring an antenna structure to controldifferential and common mode current to cause isolation between anynumber of antenna ports can be applied to any of the antenna embodimentsdescribed herein.

FIG. 31 illustrates an exemplary multiband antenna 3100 in accordancewith one or more embodiments. The antenna 3100 can include a low bandstructure comprising two low band antenna elements 3102, 3104 connectedby a connecting element 3106. A fixed or variable reactive element 3126such as a fixed or variable inductor L is provided in the connectingelement 3106 to provide control (reduction) of the mutual couplingbetween feedpoints for the low band element by varying the electricallength of the connecting element 3106 in accordance with the disclosuresof U.S. Pat. No. 7,688,273, the disclosure of which is incorporated byreference herein in its entirety. Similarly, a connecting element 3116can be provided between the high band antenna elements 3112, 3114. Afixed or variable reactive element 3136 such as a fixed or variableinductor L can be provided in the connecting element 3116 to providecontrol (reduction) of the mutual coupling between feedpoints for thelow band element by varying the electrical length of the connectingelement 3116 in accordance with the disclosures of U.S. Pat. No.7,688,273.

The high band structure comprising two high band antenna elements 3112,3114 can be connected to the low band structure at feed points f1, f2.Two filters 3142 and 3144 are provided in the high band antenna elements3112, 3114 for blocking low band frequencies, thereby isolating the highband antenna elements 3112, 3114 from the low band antenna elements3102, 3104. The filters 3142 and 3144 can be passive or programmablepass band filters. In the present illustration the filters 3142 and 3144can represent high pass filters implemented with a capacitor and/orother components to achieve desired high pass filtering characteristics.To achieve similar isolation with the low band structure, the low bandantenna elements 3102, 3104 can be configured with filters 3152, 3154 toblock high band frequencies, thereby isolating the high band antennaelements 3112, 3114 from the low band antenna elements 3102, 3104. Thefilters 3152, 3154 can be passive or programmable pass band filters. Inthe present illustration the filters 3152, 3154 can represent low passfilters implemented with reactive and passive components that achievedesired low pass filtering characteristics.

By having a structure associated with low band resonance and a separatestructure associated with high band resonance, the low band structurecan be advantageously designed or optimized independently of the highband structure and vice-versa. A further advantage is that the low bandor high band structures may separately take on different antenna designrealizations, e.g., monopole, loop, Planar Inverted “F” antenna (PIFA),etc. allowing the designer to select the best option for the electricaland mechanical design requirements. In one exemplary embodiment, the lowband structure may be a monopole, while the high band structure may be aPIFA.

A separate network is provided for each structure. The low bandstructure can use a fixed or variable inductive bridge 3126 as aninterconnecting element 3106. The high band element is fed from thecommon feedpoint, but with a high pass network 3142, 3144—the simplestbeing a series capacitor with low reactance at the high band frequenciesand higher reactance at the low band frequencies. In addition, the lowband antenna elements 3102, 3104 can be configured with variablereactive components 3122, 3124 to perform aperture tuning which enablesshifting of the low band resonance frequency of the low band structure.The reactive components 3122, 3124 can be independently controlled sothat the resonance frequency of low band antenna element 3102 can beindependently controlled from the low band resonance frequency of lowband antenna element 3104. The reactive components 3122, 3124 can berepresented by switched inductors which can be aggregated or reduced tovary the electrical length of the low band antenna elements 3102, 3104,respectively.

Similarly, the high band antenna elements 3112, 3114 can be configuredwith variable reactive components 3132, 3134 to perform aperture tuningwhich enables shifting of the high band resonance frequency of the highband structure. The reactive components 3132, 3134 can be independentlycontrolled so that the resonance frequency of high band antenna element3112 can be independently controlled from the high band resonancefrequency of high band antenna element 3114. The reactive components3132, 3134 can also be represented by switched inductors which can beaggregated or reduced to vary the electrical length of the high bandantenna elements 3112, 3114, respectively.

The aforementioned structures, enable high band tuning to be performedrelatively independent of low band tuning, providing a simpler designprocess and better performance than antennas not having such separatestructures. Other more complex networks may also be used advantageouslyto separate the interdependence of the high and low band structuresstill using a common feedpoint for a MIMO branch such as shown in FIG.31. The method illustrated in FIG. 31 is not limited to 2×2, 2×1 MIMO or2 feed antennas used for diversity applications, and may be extended tohigher branch order MIMO antennas, e.g., 3×3, etc.

A number of factors affect antenna performance in a hand held mobilecommunication device. While these factors are related, they generallyfall into one of three categories; antenna size, mutual coupling betweenmultiple antennas, and device usage models. The size of an antenna isdependent on three criteria; bandwidth of operation, frequency ofoperation, and required radiation efficiency. Bandwidth requirementshave obviously increased as they are driven by FCC frequency allocationsin the US and carrier roaming agreements around the world. Differentregions use different frequency bands, now with over 40 E-UTRA banddesignations-many overlapping but requiring world capable wirelessdevices to typically cover a frequency range from 698 to 2700 MHz.

A simple relationship exists between the bandwidth, size, and radiationefficiency for the fundamental or lowest frequency resonance of aphysically small antenna.

$\begin{matrix}{\frac{\Delta \; f}{f} \propto {\left( \frac{a}{\lambda} \right)^{3}\eta^{- 1}}} & (1)\end{matrix}$

Here a is the radius of a sphere containing the antenna and itsassociated current distribution. Since a is normalized to the operatingwavelength, the formula may be interpreted as “fractional bandwidth isproportional to the wavelength normalized modal volume”. The radiationefficiency η is included as a factor on the right side of (1),indicating that greater bandwidth, is achievable by reducing theefficiency. Radio frequency currents exist not only on the antennaelement but also on the attached conductive structure or “counterpoise”.For instance, mobile phone antennas in the 698-960 MHz bands use theentire PCB as a radiating structure so that the physical size of theantenna according to (1) is actually much larger than what appears to bethe “antenna”. The “antenna” may be considered a resonator that iselectromagnetically coupled to the PCB so that it excites currents overthe entire conductive structure or chassis. Most smartphones exhibitconductive chassis dimensions of approximately 70×130 mm, which from anelectromagnetic modal analysis predicts a fundamental mode near 1 GHzsuggesting that performance bandwidth degrades progressively at lowerexcitation frequencies. The efficiency-bandwidth trade-off is complexrequiring E-M simulation tools for accurate prediction. Results indicatethat covering 698-960 MHz (Bands 12, 13, 17, 18, 19, 20, 5 and 8) with acompletely passive antenna with desirable antenna size and geometrybecomes difficult without making sacrifices in radiation efficiency.

Factors determining the achievable radiation efficiency are not entirelyobvious, as the coupling coefficient between the “antenna” and thechassis; radiative coupling to lossy components on the PCB; dielectricabsorption in plastic housing, coupling to co-existing antennas; as wellas losses from finite resistance within the “antenna” resonatorstructure, all play a part. In most cases, the requirements imposed byoperators suggest minimum radiation efficiencies of 40-50%, so thatmeeting a minimum TRP requirement essentially requires tradeoffs betweenthe power amplifier (PA) output and the achievable antenna efficiency.In turn, poor efficiency at the antenna translates to less battery life,as the PA must compensate for the loss.

Prior to concerns over band aggregation, wireless devices operated onone band at a time with need to change when roaming. Consequently, therequired instantaneous bandwidth would be considerably less than thatrequired to address worldwide compatibility. Take a 3G example forinstance, where operation in band 5 from (824-894 MHz) compared tooperation in bands 5 plus 8 (824-960 MHz). Then, add the requirementsfor band 13 and band 17 and the comparison becomes more dramatic—824-960vs. 698-960 MHz. This becomes a problematic as legacy phone antennassupport pentaband operation but only bands 5 and band 8. Given equation(1) several choices exist. The most obvious would be to increase theantenna system size, (i.e. the antenna and phone chassis footprint)and/or to reduce the radiation efficiency. Since 4G smartphones require2 antennas, neither approach is necessarily desirable from an industrialdesign standpoint, although it is possible to cover the 700-2200 MHzbands with a completely passive antenna in a space allocation of6.5×10×60 mm.

Various alternative antenna configurations are the following: limit theantenna(s) instantaneous bandwidth within current antenna spaceallocations to allow use of 1 or more antennas without compromising theindustrial design (Antenna Supplier motivation); make the antenna(s)smaller to achieve a compact and sleek device with greater functionalityby limiting the instantaneous bandwidth with same or improved antennaefficiency (OEM motivation); improve the antenna efficiency, andtherefore the network performance by controlling the antennainstantaneous frequency/tuning (Operator motivation); make the antennaagile to adapt to different usage models (OEM/User/Operator motivation);or combinations of the above.

The simplest approach can be to limit the instantaneous operation to asingle band to satisfy the protocol requirements for a single region. Tosatisfy the roaming requirements, the antenna could be made frequencyagile on a band-by-band basis. This approach represents the most basictype of “state-tuned” antenna.

Various embodiments disclosed herein are directed to an antenna thatseparates the fundamental (low band) resonance from the high bandresonance by using two separate structures, which are connected at thefeedpoint—thus accomplishing the goal of achieving a MIMO or Diversityantenna with each feed exhibiting a multiband capability, and wherebyeach feed is optimally isolated from the opposite feed. By way ofnon-limiting example, in some implementations, high band frequencies canrange from 1710 to 2700 MHz, and low band frequencies can range from 500to 960 MHz.

The exemplary embodiments allow for tuning of the first resonance of theantenna to accommodate multiple operational bands depending on a tuningstate, and broadband operation on the high bands (e.g., 1710-2170 MHz,or 1710-2700 MHz) independent of the low band tuning state.

Referring to FIG. 32A, an example is shown that is illustrative ofsingle low band-multiple high band aggregation compatibility. The highband radiation efficiency in this case can remain essentially the sameindependent of the low band tuning state, but the low band resonancefrequency is able to be tuned in discrete frequency increments accordingto the equivalent electrical length, as selected by the seriesinductance Lvar which is shown in FIG. 32B. The variable inductance canbe created using discrete reactive elements such as inductors and aswitching mechanism such as an SP4T switch. The configuration as shownyields 3 different inductances depending on which state the switch isin: (state 1) LVAR=L3+L4+L5 (switch connects to pole 1 or 4); (state 2)LVAR=Lpath2∥L3+L4+L5 or approximately L4+L5 (switch connects to pole 2);or (state 3) LVAR=Lpath3∥(L3+L4)+L5 OR approximately L5 (switch connectsto pole 3). In this embodiment, Lpath2 and Lpath3 refer to theequivalent inductances of the circuit paths through the switch. Keepingthe inductors close to the switch can minimize or otherwise reduce thepath inductances such that the discrete inductors are essentiallyshorted out by the switch.

The antenna incorporates a main structure that has a fundamentalresonance at the lowest frequency band. The solution employs a multibandantenna having 3 low band tuning states as shown in FIG. 32B. State 1includes a low band (fundamental) resonance suited for LTE 700 (698-742MHz) operation: State 2 includes a low band resonance suited to GSM 850(824-894 MHz) operation, and state 3 a low band resonance suited to GSM900 (880-960 MHz).

The high band resonance (1710-2170 MHz) can be reasonably independent ofthe tuning state for the low band by nature of the separation of the lowand high band radiating elements from the feedpoints. The low bandtuning can be accomplished by switching different reactive components inbetween the feedpoint and the radiating structure. The high bandoperation of the antenna can be governed primarily by the auxiliaryradiating section at the terminus of the capacitor opposite thefeedpoint. The capacitor functions primarily as a high pass filter todecouple the feedpoint from the high and low bands portions of theantenna. In this way, signals at different operating bands can bedirected to the appropriate radiating section of the combined antenna.The high band resonance can be determined in part by the electricallength of the high band portion of the antenna (indicated in theillustration by horizontal conductive segments). In other embodiments,the capacitor may be a highpass, bandpass, or tunable filter. In asimilar manner, the path from the feedpoint to the low band radiatingportion of the antenna may include a low pass, bandpass or tunablefilter.

Tuning can be accomplished using a switching device such one capable ofSP4T operation. In one embodiment, a solid state silicon-based FETswitch can be used in each leg of the antenna to alter the seriesinductance presented to the antenna feedpoint, thereby lowering theresonant frequency as a function of the amount of inductance added.Although inductors are used in this embodiment, other reactivecomponents may also be used for the purpose of altering the electricallength of the low band portion of the antenna radiating structureincluding capacitive elements. The switch may be of various types suchas a mechanical MEMS type device, a voltage/current controlled variabledevice, and so forth. The switch may also be configured with multiplepoles and with any throw capability needed to select the number oftuning states required for antenna operation. The number of throws canestablish the number of tuning states possible, which in turn isdictated by the number of frequency bands to be supported. While threestates are shown in the illustrated embodiment, any number of states canbe utilized corresponding to any number of frequency bands or ranges. Inone embodiment, a pair of adjustable reactive elements (e.g., fixedinductors coupled with switching mechanisms) can be coupled withcorresponding pairs of feedpoints, and the tuning can be performed bysettings each of the adjustable reactive elements to the same tuningstate among the group of tuning states.

Referring to FIG. 33A, a separate but co-located high band element isshown with feed points F1H and F2H that allows for compatibility with RFtransceiver front end designs requiring separate low- and mid- or low-and high-band connections to the antenna. The variable inductance can becreated using discrete inductors and a SP4T switch as shown. Theconfiguration as shown yields 3 different inductances depending on whichstate the switch is in: (state 1) LVAR=L3+L4+L5 (switch connects to pole1 or 4); (state 2) LVAR=Lpath2∥L3+L4+L5 or approximately L4+L5 (switchconnects to pole 2); or (state 3) LVAR=Lpath3∥(L3+L4)+L50R approx. L5(switch connects to pole 3). Lpath2 and Lpath3 refer to the equivalentinductances of the circuit paths through the switch. Keeping theinductors close to the switch minimizes the path inductances such thatthe discrete inductors are essentially shorted out by the switch.

The exemplary antennas can provide better radiation efficiency and/orsmaller size compared to an untuned antenna by nature of the tuning toeach band of operation separately. The reactive elements (e.g.,inductors and their associated inductance) can establish the electricallength of the low band elements, and therefore can provide for adjustingthe low band resonance (tuning) Referring additionally to FIGS. 34A-35B,antenna structures that enable tuning to each band of operationseparately while also providing for desired filtering through use oflow-pass-filters and high-pass-filters as illustrated. It should benoted the embodiments for aperture tuning shown in FIGS. 32B, 33B, 34Band 35B can replaced with other suitable embodiments such as those shownin FIGS. 32C, 33C, 34C and 35C.

Further, the fundamental mode associated of the antenna low bandresonance can be tuned by adjustment of the electrical length of the lowband portion of the antenna via reactive elements which may exhibiteither inductive or capacitive characteristics. As illustrated in FIG.32A, discrete inductors are shown in a series connection between theantenna feed points and the radiating element end plates on the eachside of the antenna, thereby increasing the equivalent electricallength. The use of separate or discrete components is intended to beillustrative of the principle, but by no means limiting to scope of thesubject disclosure. In one or more embodiments, the techniques and/orcomponents of the exemplary embodiments described herein that providefor antenna tuning can be utilized in conjunction with techniques and/orcomponents described with respect to U.S. Pat. No. 7,688,273. In one ormore embodiments, the techniques and/or components of the exemplaryembodiments described herein that provide for antenna tuning can beutilized in conjunction with techniques and/or components described withrespect to U.S. patent application Ser. No. 14/285,262, entitled“Antenna Structures and Methods Thereof”, filed on May 22, 2014, underattorney docket number 5000-0192 (2013-03B).

FIG. 36A depicts an illustrative embodiment of a near field sensor 3620utilized with other RF components of a communication device 3600. In oneembodiment near field sensor 3620 can comprise a first log detector 3622that receives signals from a directional coupler 3604, a second logdetector 3626 that receives signals from a near field probe 3624. Inanother embodiment, such as shown in FIG. 36B, the near field sensor3620 can comprise a single log detector 3626 that receives signals fromone or more near field probes 3624 selectable by a switch 3625. Insteadof relying on a directional coupler 3604 such as shown in FIG. 36A, thecontroller 3632 of FIG. 36B can be configured to determine a magnitudeof a forward feed signal supplied by the transmitter 3602 based on aknown state of the transmitter 3602, which can be determined from alook-up table. That is, the controller 3632 may be aware that thetransmitter 3602 has been configured to transmit at a desired magnitudeas a result of configuring components of the transmitter 3602 such as anamplifier or otherwise. The controller 3632 can utilize a look-up tablecomprising magnitudes indexed according to known states of thetransmitter 3602, or can determine the magnitude of a transmit signalbased on an algorithm implemented by the controller 3632. The look-uptable can be created by characterizing the transmitter 3602 in a lab ormanufacturing setting. Accordingly, the embodiments that follow can bebased on either the embodiments of FIG. 36A or 36B.

The near field probe 3624 can be a small trace of metal serving as aminiature antenna that can receive radiated energy from an adaptiveantenna 3610 and which can have a very small parasitic effect (if any)on the adaptive antenna 3610, thereby unaffecting the original operatingcharacteristics of the adaptive antenna 3610. The near field probe 3624can be located on a printed circuit board (PCB), a housing assemblycomponent or some other suitable location of the communication device3600 that enables placement of the near field probe 3624 at a particularperspective of the adaptive antenna 3610. As depicted in FIGS. 36A and36B, more than on near field probe 3624 can be used at various locationof the communication device 3600 as will be described below.

The output signals of the first and second log detectors 3622 and 3626can be supplied to a difference circuit 3628 that produces a differencesignal supplied to an analog to digital converter (A2D) 3630, which inturn supplies a digital value to a controller 3632 for processing. Thedifference signal represents a difference in magnitude between thesignals supplied by the first and second log detectors 3622 and 3626.The signal supplied by the first log detector 3622 can represent ameasure of a forward feed signal supplied by an RF transmitter 3602. Thedifference in magnitude between the forward feed signal and the signalmeasured by the near field probe 3624 can be used to detect a change ina resonance frequency of the adaptive antenna 3610. In the embodiment ofFIG. 36B, the controller 3632 would be supplied a digital value from theA2D 3630 that corresponds only to the magnitude of the near field signalgenerated by the log detector 3626. The controller 3632 in turn canobtain a digital representation of the magnitude of the forward feed (ortransmit) signal supplied by the transmitter 3602 by retrieving amagnitude value from a look-up table based on a known state of thetransmitter 3602 or it can obtain it by way of a computational approachthat determines the magnitude from the known state of the transmitter3602. Once the forward feed magnitude is determined, the controller 3632can determine a difference between the forward feed and near fieldsignal magnitudes using digital signal processing techniques, andthereby detect a change in a resonance frequency of the adaptive antenna3610.

The adaptive antenna 3610 can be an antenna structure such as any of theembodiments described above. For instance, the adaptive antenna 3610 canbe represented by one of the embodiments of FIGS. 31-35, which enableprogramming of the resonant frequency of the antenna. If the detectedchange of the resonance frequency of the adaptive antenna 3610 isdetermined to be undesirable based on the difference signals supplied bythe A2D 3630, the adaptive antenna 3610 can be tuned by the controller3632 by changing the electrical length of the antenna by reconfiguringthe structure of the antenna (e.g., adding or removing antennacomponents), by adding or removing reactive components such as theembodiments of FIGS. 32B, 33B, 34B and 35B, or changing the reactance ofa component of the antenna (e.g., a variable capacitor or variableinductor). As the adaptive antenna 3610 is tuned by the controller 3632,the controller 3632 can also be configured to program a matching network3608 or supply the tuning data of the adaptive antenna 3610 to aseparate controller of the matching network to enable programming of thematch network to adapt to changes applied to the adaptive antenna 3610while tuning its resonant frequency. It is noted that the near fieldsensor 3620 as described herein can be used with other antennastructures not described in the subject disclosure.

What follows is an illustrative algorithm for tuning the adaptiveantenna 3610 based on near field RF power measurements. We begin byassuming the adaptive antenna 3610 is tunable with a total of N states.The adaptive antenna 3610 can be set to a particular state, k, based onoperating conditions such as a band of operation. The A2D 3630 providesthe controller 3632 a relative power measurement, Y, at a specifictransmit frequency in use. In one embodiment, an objective is to improveY. To do this, the k is stepped at time interval, τ, and the response ofY is used to determine if the state should be incremented, decremented,or kept the same. The time τ may be chosen to be longer than a responsetime of other power control loops used by the communication device 3600and/or other network operations that may be dependent on the transmitpower level from the communication device 3600. With this in mind, atuning algorithm can be described as follows:

Let k0 be the nominal setting for the state value

Let the value of k alternate between k0, k0+1 and k0−1:

k(nτ)=k0+cos(nπ/2), where n is the number of time increments

For each state, measure the probe response Y(nτ)

Calculate the slope, m=dY/dk

Calculate the running average mbar, over M time intervals,

m=Σ_(i=n-M) ^(n)m(nτ)

Compare mbar to a threshold τ to determine whether to change the state

If mbar≦δk=k−1

If mbar≦δk=k+1

If |mbar|<δ k is unchanged

In an embodiment where multiple near field probes 3624 are used, theabove algorithm can be changed so that delta measurements of Y areaveraged over time. Averaging delta Y readings can increase thereliability of the measurement. In one embodiment, the controller 3632can select an individual near field probe 3624 using a programmablemultiplexer (S) 3625. The controller 3632 can then apply the abovealgorithm using only those near field probes 3624 that provide a delta Ywith desirable results. This approach can be applied to each near fieldprobe 3624 until one or more near field probes 3624 are identified asproviding desirable tuning results of the adaptive antenna 3610. Thecontroller 3632 can then average the delta Y magnitudes measured foreach near field probe 3624, or select only one of the identified nearfield probes 3624 for tuning the adaptive antenna 3610. In amulti-antenna system where the transmitter 3602 can transmit from anyone of a plurality of adaptive antennas 3610, the near field sensor 3620can be configured to choose a near field probe 3624 closest to theantenna used during transmission for performing the above algorithm. Inanother embodiment, if for a selected near field probe 3624 the nearfield sensor 3620 is not performing due to unrepresentative informationsupplied by the selected near field probe 3624 (e.g., a user's finger ischanging the performance of the selected near field probe 3624) then thecontroller 3632 can be adapted to avoid the identified near field probe3624 and perform averaging from other probes unaffected by theenvironment effect.

In other embodiments averaging can be performed for more than one nearfield probe 3624 associated with a single adaptive antenna. In yetanother embodiment, multiple near field probes 3624 can be placed oneach radiating element of an antenna (high band, mid band, and low bandnear field probes). In another embodiment, one near field probe can beused per radiating element of an antenna. In yet another embodiment, apad of an integrated circuit (IC) can be used as a near field probe 3624of an antenna.

Multiple near field probes 3624 can be used at different locations ofthe communication device 3600 using a multiplexer 3625 to select betweennear field probes 3624 to get a measurement of radiated energy of theadaptive antenna 3610 from different perspectives (e.g., bottom, top,sides). FIG. 37 provides a placement illustration of six near fieldprobes 3624 on a printed circuit board. The near field probes 3624 ofFIG. 37 are sized to yield −30 dB of coupling with the antenna structureshown. Low band simulations were performed on an antenna structure witha PCB ground of 110 mm×60 mm. FIG. 38 depicts six use cases applied tothe circuit board of FIG. 37. The gray sections of each use caserepresent body tissue or steel. The simulation further assumed a 2 mmair gap between a furthest model element and a log material.

FIG. 39 depicts return loss plots (S11) for steel and body tissue andtheir respective efficiencies for each use case. Compared with freespace, the resonant frequency of the antenna shifts from free space from−150 to +60 MHz. The efficiency changes from −18 to −2 dB. The Smithcharts of FIG. 40 depict steel and body tissue plots of all probes forall use cases. FIG. 41 depicts magnitude and phase response of probe 2which is closest to an end of the antenna. FIG. 42 depicts the magnitudeand phase response of probe 4 which is to the left of the antenna. Inthis latter embodiment, the response is not well behaved, whichillustrates that not all probe locations may provide optimal results.FIG. 43 depicts phase shift versus antenna frequency shift plots forprobe 2. These plots show that the phase shift experienced at probe 2correlates with an observed frequency shift. The sensitivity is about0.24 degrees per MHz. FIG. 44 depicts power shift versus antennaradiated power shift plots for probe 2. These plots show that themagnitude shift experienced at probe 2 correlates worst with an observedpower shift. Probe 7 (opposite end of ground) correlates best with theradiated power shift. The sensitivity is about 0.28 degrees per radiatedpower shift.

FIGS. 45-48 depict simulations for an aperture tuned antenna model. FIG.45 depicts two embodiments having near field probes placed near and farfrom an antenna located at one end of a PCB having dimensions of 60mm×126 mm. The antenna is designed for a low band nominal frequency of860 MHz. A variable inductor (Ltune) can be changed to shift theresonant frequency of the antenna. The models shown in FIG. 45 can beused to determine how much improvement can be made by changing Ltune inseveral loading scenarios.

FIG. 46 depicts free space resonance tuning using Ltune. Return loss(S11) is presented in the pots of FIG. 46 as a function of the value ofLtune. The simulations assume in free space a nominal setting of L=5.6nH at a resonant frequency of 860 MHz. The simulation methodologyincluded running each of the models of FIG. 45 with all of the loadingcases shown in FIG. 38. The value of Ltune is then swept for eachloading case. A determination is then made of the maximum radiated powerat the desired operating frequency (fc) over the sweep of Ltune. Themaximum radiated power is then compared based on metrics such asmaximizing a magnitude of S21 (power coupled to the near field probe),minimizing a magnitude of S11, and normalizing S21 phase to 0 degrees(the normalized phase is the measured S21 phase minus the referencephase delay at resonance center determined from the free space model orby the actual phase when the antenna is tuned to the desired frequency).

FIG. 47 depicts an efficiency gain by optimization of tuning for thecase of the near field probe nearest the antenna in FIG. 45. FIG. 47shows an improvement in radiated power from 0 to 2.2 dB depending onloading case. It is further observed that an ideal optimization of S21or S11 results in nearly the same gain as optimizing on radiated poweritself. It is also observed that optimizing on S21 phase also works wellexcept in five of the six loading cases. FIG. 48 depicts an efficiencygain by optimization of tuning for the case of the probe opposite theantenna in FIG. 45. FIG. 48 shows again that optimization of S21 or S11are reliable. However, in this case, optimizing on S21 phase workspoorly in the situation where near field probe is far from the antenna.

FIG. 49 depicts an illustrative implementation of a near field sensor.In one embodiment, the near field sensor can comprise a transmissionline coupled to a directional coupler, an RF IC (e.g., an Analog DevicesAD8302) powered by a battery for measuring amplitude and phase between aforward feed signal supplied by the directional coupler, and a signalsupplied by the near field probe. The magnitude and phase measured bythe RF IC can be supplied to a controller that performs algorithmicsteps as described earlier for detecting a change in a desire frequencyof the antenna. RF paths AB and AC can be configured to 90 degrees ofpath difference for nominal free space, which is the center of themeasurement range for phase of the RF IC. RF paths AB and AC can be setto ˜30 dB path loss, the center of the input range being −30 dBm. FIG.50 depicts a PCB layout of the near field sensor.

It is noted that any of the adaptive antenna embodiments described inthe subject disclosure can be applied to multiple MIMO configuration(2×2, 4×4, etc.) or diversity configurations. It is further noted thatadaptive antenna embodiments described in the subject disclosure can beapplied to multiband antenna structures. In such configurations, thesubject disclosure enables support for carrier aggregation of multiplebands for simultaneous transmission or reception in MIMO or diversityconfigurations, while maintaining at least some isolation betweenantenna ports.

FIG. 51 depicts an illustrative embodiment of a method 5100 that can beapplied to the embodiments of the subject disclosure. The method 5100can begin at step 5102 where circuitry (such as a log detector) can beused to measure a magnitude of a forward feed signal supplied by adirectional coupler coupled to an adaptive antenna by way of a matchingnetwork. At step 5104 a near field probe can be used to measure amagnitude of radiated energy from the adaptive antenna. At step 5106 adifference between the magnitude of the forward feed signal and theradiated energy of the antenna can be determined. At step 5110 adetermination can be made whether the difference in magnitude results ina change in the operating frequency (fc) of the adaptive antenna. Ifthere is no change (or an insignificant change) in the operatingfrequency (fc) is detected, method 5100 can be adapted for repeatingsubsequent iterations of method 5100 beginning from step 5102. If thechange in the operating frequency (fc) is considered significant, atstep 5112 a frequency offset error can be determined according to thedifference in magnitude determined at step 5106.

If the frequency offset error is considered insignificant, method 5100can be repeated in subsequent iterations beginning from step 5102. If,however, the frequency offset error is considered undesirable (orunacceptable), at step 5114 the adaptive antenna can be tuned by, forexample, varying the electrical length of the antenna based on an Ltunevalue calculated from the difference in magnitude between the forwardfeed signal and the radiated energy of the antenna. The Ltune value canbe used to configure a switched array of inductors such as shown inFIGS. 32B, 33B, 34B and 35B. Alternatively, a tuning value can becalculated from the difference in magnitude between the forward feedsignal and the radiated energy of the antenna, which can be used to tuneone or more programmable variable reactive elements (such as one or morevariable capacitors, one or more variable inductors, or combinationsthereof) coupled to the antenna that reconfigure the electrical lengthof the antenna, and thereby change the operating frequency of theantenna to a desirable frequency.

FIG. 52 depicts another illustrative embodiment of a near field sensor5210. In this embodiment, the near field sensor 5210 can be configuredto perform operations including measuring phase changes in signalsradiated by the adaptive antenna 3610 using one or more near fieldprobes 3624, measuring a phase of a forward feed signal using adirectional coupler 3604, determining a phase offset between the phaseradiated energy emitted by the adaptive antenna 3610 and the phase ofthe forward feed signal supplied to the adaptive antenna 3610 using aphase detector 5212, and tuning the operating frequency of the adaptiveantenna 3610 according to the phase offset using the controller 3632.FIG. 53 depicts a phase detector from Analog Devices™ (AD8302), whichcan be used in the subject disclosure. The phase detector of FIG. 53 canbe used for measuring gain/loss and phase up to a frequency of 2.7 GHz.The phase detector includes dual demodulating log amps with a phasedetector input range of −60 dBm to 0 dBm in a 50 Ohm system.

In one embodiment, an inductive, capacitive, and resistive (LRC) model(such as shown by reference 5214) of an unloaded adaptive antenna 3610can be characterized by an antenna supplier. The LRC characteristics ofthe adaptive antenna 3610 can be characterized at a time of manufactureof the adaptive antenna 3610. Characterization and/or factorymeasurements can be stored in a look-up table (or hardcoded) in analgorithm executed by a controller or ASIC design for calculating afrequency offset measurement to perform tuning of the adaptive antenna3610. A phase calibration measurement can also be performed on adetector chain (directional coupler 3604, phase detector 5212, nearfield antennas 3624) to remove phase measurement error in the algorithm.Phase calibration may depend on frequency of operation of the adaptiveantenna 3610. Accordingly, a phase calibration look-up table can beimplemented that is frequency dependent to accommodate changes incalibration based on frequency of operation of the adaptive antenna3610.

In one embodiment, the impedance of the LRC model 5214 can be describedby the following equation:

$Z = {R + {j\; \omega \; L} + \frac{1}{j\; \omega \; C}}$

where ω=2πf and f is frequency. The complex power delivered can bedescribed by the equation:

$S = {\frac{V^{2}}{R + {j\; \omega \; L} + \frac{1}{j\; \omega \; C}} = {\frac{V^{2}}{R + {j\left( {{\omega \; L} - \frac{1}{\omega \; C}} \right)}} = {\frac{R - {j\left( {{\omega \; L} - \frac{1}{\omega \; C}} \right)}}{R^{2} + \left( {{\omega \; L} - \frac{1}{\omega \; C}} \right)^{2}}*V^{2}}}}$

Based on the above equation, phase of S can be determined according toequation:

$\varnothing = {{atan}\left( \frac{\frac{1}{\omega \; C} - {\omega \; L}}{R} \right)}$

When Ø=0, the antenna model 5214 is at resonance.

Solving for ω:

${{\omega^{2}L} + {R*\tan \; \varnothing*\omega} - \frac{1}{C}} = 0$${\omega (\varnothing)} = \frac{{{- R}*\tan \; \varnothing} + \sqrt{{R^{2}*\tan^{2}\varnothing} + {4\; {L/C}}}}{2\; L}$${f(\varnothing)} = \frac{\omega (\varnothing)}{2\; \pi}$

From the above equations, the following conditions can be assessed:

-   -   If f_(operation (op)=f) _(resonance (r)), then Ø=0, and        Ω=1/√{square root over (LC)}=ω₀    -   If f_(op)<f_(r), then Ø>0, and ω<1/√{square root over (LE)}    -   If f_(op)>f_(r), then Ø<0

Based on the above observations, one embodiment of an algorithm fordetecting a change in an operating frequency of the adaptive antenna3610 can comprise the following steps:

-   -   Phase detector 5212 measures a differential phase Ø    -   The controller 3632 can then calculate a measured frequency        according to ω_(measured)=f(Ø)    -   The controller 3632 knows the desired frequency ω_(target)    -   The controller 3632 can then calculate a relative scaling error        in the measured versus the target frequency        ∝_(offset)=ω_(measured)/ω_(target)    -   As noted earlier, a phase calibration measurement can be        performed on the detector chain to remove phase measurement        error. This can be factored into calculating the ∝_(offset)    -   ∝_(offset) may be integrated or averaged over a time interval or        a number of samples to reduce noise or sensitivity to transients        shorter than desired time interval    -   Also described earlier, the antenna resonance can be modeled        according to ω₀=1/√{square root over (LE)}    -   To correct a detected offset in frequency back to a target        frequency (ω_(target)), target), the present operating frequency        (ω_(measured)) can be multiplied by 1/∝_(offset)    -   Removing the offset can be accomplished by tuning L or C of the        antenna. As noted earlier, the LC modeling of the antenna can be        characterized in a lab setting, during manufacturing, or by        other suitable means. An aperture tuner using a switch array of        inductors such as shown in FIGS. 32B-35B can be used to tune the        adaptive antenna 3610. Tunable capacitive devices can also be        used in place of the tunable inductors or combinations thereof    -   To remove the detected frequency offset, the adaptive antenna        3610 can be tuned such that the product of LC is changed by a        factor of ∝² _(offset) to return ω_(measured) to ω_(target).    -   The plots of FIG. 54 depict phase error versus frequency and        power level of the phase detector (AD8302 shown in FIG. 53). To        improve the accuracy of phase detection data provided by the        phase detector to the controller, the phase detector can be        utilized in a configuration whereby a phase delay from a near        field probe is set using a length of delay line such that an RF        signal reaching the phase detector input from the near field        probe is approximately 90 degrees out of phase with an RF signal        reaching another the phase detector input from the directional        coupler under nominal conditions, taken as a free space        condition. In another embodiment, the delay line can be instead        applied to the RF signal supplied by the directional coupler.

FIG. 55 depicts an illustrative embodiment of a method 5500 forfrequency offset detection and tuning based on phase measurements.Method 5500 can begin with step 5502 in which a forward feed signal(e.g., a transmit signal supplied by the transmitter 3602) is measuredby way of the directional coupler 3604 and supplied to the phasedetector 5212. Similarly, at step 5504, the near field probe 3624 cansupply a measure of a signal representing radiated energy from theadaptive antenna 3610. As noted in other embodiments, a switch, such asshown in FIG. 52, can be used to select from a plurality of near fieldprobes 3624 to provide measurements of radiated energy from the adaptiveantenna 3610 from multiple positions of the probes. The controller 3632can selectively use multiple measurements from one or more near fieldprobes 3624 to improve the accuracy of the algorithm.

At step 5506, the phase detector 5212 can generate a phase differentialbased on the phase of the signals measured at steps 5502 and 5504. Adigital representation of the phase differential is then supplied by theA2D 3630 to the controller 3632 for calculating a frequency offset(∝_(offset)). The controller 3632 at step 5505 can retrieve phasecalibration data from a look-up table based on a target frequency ofoperation or a measured frequency of operation. In one embodiment thelook-up table can further depend on an open loop state of the antennaand use case information such as whether a communication deviceutilizing the antenna structure of the subject disclosure is being heldby a user's hand, whether the communication device is in a position nextto a user's ear, and so on. The phase calibration data can be used toadjust calibration errors in the phase differential. To calculate afrequency offset as described above, the controller 3632 can alsoretrieve from a look-up table at step 5507 an LRC model of the adaptiveantenna 3610, which may also be frequency dependent. The calibrationdata and the LRC model can be stored in a memory of a communicationdevice that integrates the embodiments of FIG. 52. At step 5510, thecontroller 3632 can determine whether a change in frequency from adesired target frequency has occurred. A change in frequency can bedetermined when, for example, ∝_(offset) is greater than or less thanunity (i.e., 1).

At step 5512, the controller 3632 can be configured to make adetermination whether the frequency offset is significant enough towarrant retuning of the measured operating frequency of the adaptiveantenna 3610. To avoid excessive retuning of the adaptive antenna 3610,the controller 3632 can be configured to compare ∝_(offset) to a rangeof thresholds. For example, when ∝_(offset)>1, ∝_(offset) can becompared to a first threshold. The first threshold can represent anacceptable frequency overshoot range of the frequency measured(w_(measured)) above the target frequency (w_(target)). Similarly, when∝_(offset)<1, ∝_(offset) can be compared to a second threshold. Thesecond threshold can represent an acceptable frequency undershoot rangeof the frequency measured (w_(measured)) below the target frequency(w_(target)). The first and second thresholds can be based onspecifications provided by a network provider. Thus, when the frequencymeasured exceeds the first threshold or is below the second threshold,the controller 3632 can proceed to step 5514 where it retunes theadaptive antenna 3610 as previously described in the subject disclosureto bring ∝_(offset) closer to unity.

It is further noted that more than two thresholds (e.g., two thresholdsfor detecting large offsets and two additional thresholds for detectingsmaller offsets) can be used to enable a determination when a coarsetuning versus a fine tuning of the adaptive antenna 3610 is required.

In another embodiment of the subject disclosure, method 5500 of FIG. 55can be adapted to use a first look-up table that maps a detected phaseoutput voltage to a frequency shift, and a second lookup table that mapsantenna tuning states to the frequency shift determined from the firstlook-up table. Accordingly, when the controller 3632 reads a voltagesignal from the phase detector 5212 in digital format provided by theA2D 3630, the controller 3632 proceeds to look up a frequency shift inthe first look-up table, then utilizes the frequency shift to indexthrough the second look-up table to obtain one or more state incrementsto shift the frequency of the antenna to a desired state. The one ormore state increments can represent an adjustment of the electricallength of the adaptive antenna 3610 utilizing an array of selectableinductors such as shown in FIGS. 32B-35B.

In also noted that the methods depicted in FIGS. 51 and 55 can becombined to further enhance frequency offset detection and retuning. Forexample, in one embodiment delta averages of Y can be compared to deltaaverages of ∝_(offset), and such comparisons can be used to estimate afrequency offset. In another embodiment, delta averages of Y can becorrelated to expected delta averages of ∝_(offset), which can berecorded in a look-up table. Similarly, delta averages of ∝_(offset) canbe correlated to expected delta averages of Y, which can be recorded ina look-up table. The look-up tables can in turn be used for validationof a frequency offset calculation. Other suitable techniques can be usedto compare the results of one algorithm against the other for validationof measurements.

RF magnitude detection (embodiment of FIG. 51) and RF phase detection(embodiment of FIG. 54) can be used together to provide improved antennafrequency tuning. The RF magnitude detection provides a directmeasurement of radiated power. Phase detection yields an instantaneousestimate of the amount of frequency error, and hence the amount ofcorrection needed. In one embodiment the frequency error estimatedetermined from phase detection may be used to calculate a specificamount of frequency shift and direction for the next iterative step inthe magnitude detection algorithm. Thus phase detection algorithm ofFIG. 54 can enable the RF magnitude algorithm of FIG. 51 to close in ona desired frequency of operation.

In another embodiment, an iterative algorithm for mitigating a frequencyoffset can utilize both magnitude and phase readings as described in thesubject disclosure to determine an error estimate. The error estimatecan be based on a weighted combination of both magnitude (mbar) andphase (∝_(offset)) offsets, with the algorithm seeking to reduce thecombined error estimate instead of relying on either the magnitude orphase error estimates individually.

FIG. 56 depicts an illustrative embodiment of a system 5600 having areactance sensor 5608 for measuring a reactive load applied to theantenna 3610. The reactive sensor 5608 can be used, for example, tomeasure a load capacitance of the antenna 3610 utilizing a delta-sigmacapacitance meter or other suitable circuitry operating at a samplingfrequency fs or at some band of frequencies separate from the operatingfrequency band of the RF and/or antenna system. The reactive sensor 5608in turn supplies the controller 3632 a digital signal representing themeasured load capacitance of the antenna 3610. An RF choke 5606 can beused to restrict the flow of RF signals to the reactive sensor 5608except RF signals at or below the sampling frequency f_(s) of thereactive sensor 5608. The separation of the RF front end (i.e., 3601and/or 3602) from the reactive sensor 5608 may also be accomplished withfilters or other signal separation means, including a diplexer, aduplexer, a multiplexer or other suitable circuitry. In thisillustration, the carrier frequency f_(c) of signals received by thereceiver 3601 from the antenna 3610 or supplied by the transmitter 3602to the antenna 3610 are at a much higher frequency than the samplingfrequency f_(s) of the reactive sensor 5608, thereby enabling the RFchoke 5606 to prevent such signals from being leaked into the reactivesensor 5608. A transient suppressor 5604 can be used for ESD protection,and can utilize a diode circuit to prevent the sampling signal of thereactive sensor 5608 from being dissipated to ground. It should be notedthat loading conditions contributed by the RF choke 5606, the transientsuppressor 5604 and the matching network 3608 any other circuits ofsystem 5600 can be pre-measured at various frequencies in a controlledsetting (e.g., a lab and/or manufacturing) based on known loadingconditions of the antenna 3610. Such loading measurements can betabulated in a look-up table at various operating frequencies andutilized to improve the accuracy of capacitive load readings of theantenna 3610 by removing known errors from measurements made by thereactive sensor 5608.

In one embodiment, frequency tuning using the system 5600 of FIG. 56 canbe performed by measuring a change in reactance (e.g., capacitance) ofthe adaptive antenna 3610, converting the measured change in reactanceto a frequency offset, and tuning an operating frequency of the adaptiveantenna 3610 according to the frequency offset. An LRC model of anunloaded antenna 5602 such as shown in FIG. 56 (without Z_(IN) andC_(L)) can be characterized in a lab setting, during manufacturing, orby an antenna supplier. Characterizations of the LRC model at variousfrequencies can be stored in a look-up table (or hardcoded) whenperforming frequency tuning as will be described below.

In one embodiment, the impedance of the LRC model 5602 can be describedby the following equation, which was earlier described:

$Z = {R + {j\; \omega \; L} + \frac{1}{j\; \omega \; C}}$

where ω=2πf and f is frequency, R=R_(rad)+R_(internal) (radiationresistance+resistance of antenna), L is the equivalent inductance of theantenna, and C is the equivalent capacitance of the antenna 3610. Theabove equation can be rewritten as:

$Z = {{j\left\lbrack {{\omega \; L} - \frac{1}{\omega \; C}} \right\rbrack} + R}$

Resonance occurs when ωL=1/ωC or ω²=1/LC. It follows therefore that theresonance frequency f_(r)=½π√{square root over (LC)}.

Any external loading of the adaptive antenna 3610 may be assumed to be(at first order) a parallel capacitance C_(L). The load capacitance mayalso consist of other reactive components associated with the circuitimplementation, as indicated previously. The resonance frequency canthus be determined from f_(r)=½π√{square root over (LC_(T))}, whereC_(T)=C+C_(L). By measuring C_(L), and knowing LRC values from a look-uptable, a change in resonant frequency can be calculated.

Considering the impedance Z_(IN) of the matching circuit 3608, which maybe programmable, the impedance of the 5602 model can be furthercharacterized as impedance Z_(A), which can represent an impedancelooking in from a voltage source of the LRC model 5602. Z_(A) can bedescribed by the following equations:

$Z_{A} = {r + {j\; b} + {j\; \omega \; L} - {\frac{j}{\omega}*\left\lbrack \frac{1}{C + {CL}} \right\rbrack} + R}$$Z_{A} = {r + R + {j*\left( {b + {\omega \; L} - {\frac{1}{\omega}\left\lbrack \frac{1}{C + {CL}} \right\rbrack}} \right)}}$

where Z_(IN)=r+jb, and where r is the real impedance and b is thereactance of the matching network 3608 (whether fixed or programmable),each which can be known to the controller 3632. Resonance occurs whenthe imaginary quantity becomes zero, as described by the followingequation:

${b + {\omega \; L}} = {\frac{1}{\omega}\left\lbrack \frac{1}{C + {CL}} \right\rbrack}$

which can be rewritten in quadratic form as:

${\omega^{2} + {{b/L}*\omega} - {\frac{1}{L}\left\lbrack \frac{1}{C + {CL}} \right\rbrack}} = 0$

The variable ω can be solved for with the quadratic formula based onknown values for r, b, L, C and C_(L) (which can be measured by thereactive sensor 5606 as shown in FIG. 56). Once ω is solved, the newresonant frequency can be determined from f_(r′)=ω/2π, which can becompared to an expected resonant frequency (i.e., expected operatingfrequency of the antenna 3610) to determine a frequency offset. Knowingthe frequency offset, the controller 3632 can be programmed to adjustthe resonant frequency of the antenna by programming an aperture tunerof the antenna 3610 (such those shown in FIGS. 32B-32C, 33B-33C,34B-34C, and 35B-35C). It is noted that the reactive sensor 5608 canalso be used to measure other parameters such as inductance, oradmittance.

FIG. 57 depicts an illustrative embodiment of a third method 5700 thatcan be applied to the subject disclosure. Method 5700 can begin at step5702 where the reactive sensor 5608 measures a load capacitance of theantenna 3610. As noted earlier, the controller 3632 can retrievecapacitive offset values from a look-up table based on a known operatingfrequency of the system 5600 to remove sensing errors when measuring thecapacitive load of the antenna 3610. Based on known tabulated values ofthe LRC model of the antenna 3610 at different operating frequencies,the controller 3632 can retrieve such values based on an expectedoperating frequency to determine at step 5706 a frequency offset of theantenna 3610 (as described above) according to the load capacitancemeasured at step 5702. If a frequency offset is detected at step 5708,the controller 3632 can proceed to step 5710 to assess if the detectederror is nominal and thus can be ignored, or if it significant enough towarrant a retuning of the resonant frequency of the antenna 3610 at step5712 utilizing the aperture tuning techniques described in the subjectdisclosure. The error determination can be based on one or morethresholds applied to step 5512.

The methods depicted in FIGS. 51, 55 and 57 can be combined to improvethe accuracy of detecting a frequency offset of the antenna 3610 andmitigating such offsets through aperture tuning. For example, in oneembodiment delta averages of Y determined by method 5100 can be comparedto delta averages of ∝_(offset) determined by method 5500, and to deltaaverages of frequency offsets determined by method 5700 (which will bereferred to as Ø_(offset)) and such comparisons can be used to estimatea collective frequency offset. In another embodiment, delta averages ofY can be correlated to expected delta averages of ∝_(offset), which canbe recorded in a look-up table. Similarly, delta averages of ∝_(offset)can be correlated to expected delta averages of Y, which can be recordedin a look-up table. Similarly, delta averages of ∝_(offset) can becorrelated to expected delta averages of Ø_(offset), which can berecorded in a look-up table. Similarly, delta averages of Ø_(offset) canbe correlated to expected delta averages of ∝_(offset), which can berecorded in a look-up table. Additional combinations of these look-uptables can be created to account for all possible combinations or asubset thereof. The collection of look-up tables in turn can be used forvalidation of a frequency offset calculation by the controller 3632 byany one of the algorithms described in the subject disclosure. Othersuitable techniques can be used to compare the results of one algorithmagainst another for validation of measurements.

It is further noted that the embodiments of the subject disclosure canbe applied to mobile or stationary communication devices. Mobilecommunication devices can include without limitation cellular phones,smartphones, tablets, laptop computers, and so on. Stationarycommunication devices can include base stations such as a cellular basestation, a femto cell, a wireless fidelity access point, a small cell, amicro cell, and so on.

FIG. 58 depicts an illustrative embodiment of a communication device5800. The communication device 5800 can comprise a wireline and/orwireless transceiver 5802 (herein transceiver 5802), a user interface(UI) 5804, a power supply 5814, a location receiver 5816, a motionsensor 5818, an orientation sensor 5820, and a controller 5806 formanaging operations thereof. The transceiver 5802 can supportshort-range or long-range wireless access technologies such asBluetooth, ZigBee, WiFi, DECT, or cellular communication technologies,just to mention a few. Cellular technologies can include, for example,CDMA-1X, UMTS/HSDPA, GSM/GPRS, TDMA/EDGE, EV/DO, WiMAX, SDR, LTE, aswell as other next generation wireless communication technologies asthey arise. The transceiver 5802 can also be adapted to supportcircuit-switched wireline access technologies (such as PSTN),packet-switched wireline access technologies (such as TCP/IP, VoIP,etc.), and combinations thereof. The transceiver 5802 can be adapted toutilize any of the aforementioned adaptive antenna embodiments describedabove singly or in any combination.

The UI 5804 can include a depressible or touch-sensitive keypad 5808with a navigation mechanism such as a roller ball, a joystick, a mouse,or a navigation disk for manipulating operations of the communicationdevice 5800. The keypad 5808 can be an integral part of a housingassembly of the communication device 5800 or an independent deviceoperably coupled thereto by a tethered wireline interface (such as a USBcable) or a wireless interface supporting for example Bluetooth. Thekeypad 5808 can represent a numeric keypad commonly used by phones,and/or a QWERTY keypad with alphanumeric keys. The UI 5804 can furtherinclude a display 5810 such as monochrome or color LCD (Liquid CrystalDisplay), OLED (Organic Light Emitting Diode) or other suitable displaytechnology for conveying images to an end user of the communicationdevice 5800. In an embodiment where the display 5810 is touch-sensitive,a portion or all of the keypad 5808 can be presented by way of thedisplay 5810 with navigation features.

The display 5810 can use touch screen technology to also serve as a userinterface for detecting user input. As a touch screen display, thecommunication device 5800 can be adapted to present a user interfacewith graphical user interface (GUI) elements that can be selected by auser with a touch of a finger. The touch screen display 5810 can beequipped with capacitive, resistive or other forms of sensing technologyto detect how much surface area of a user's finger has been placed on aportion of the touch screen display. This sensing information can beused to control the manipulation of the GUI elements or other functionsof the user interface. The display 5810 can be an integral part of thehousing assembly of the communication device 5800 or an independentdevice communicatively coupled thereto by a tethered wireline interface(such as a cable) or a wireless interface.

The UI 5804 can also include an audio system 5812 that utilizes audiotechnology for conveying low volume audio (such as audio heard inproximity of a human ear) and high volume audio (such as speakerphonefor hands free operation). The audio system 5812 can further include amicrophone for receiving audible signals of an end user. The audiosystem 5812 can also be used for voice recognition applications. The UI5804 can further include an image sensor 5813 such as a charged coupleddevice (CCD) camera for capturing still or moving images.

The power supply 5814 can utilize common power management technologiessuch as replaceable and rechargeable batteries, supply regulationtechnologies, and/or charging system technologies for supplying energyto the components of the communication device 5800 to facilitatelong-range or short-range portable applications. Alternatively, or incombination, the charging system can utilize external power sources suchas DC power supplied over a physical interface such as a USB port orother suitable tethering technologies.

The location receiver 5816 can utilize location technology such as aglobal positioning system (GPS) receiver capable of assisted GPS foridentifying a location of the communication device 5800 based on signalsgenerated by a constellation of GPS satellites, which can be used forfacilitating location services such as navigation. The motion sensor5818 can utilize motion sensing technology such as an accelerometer, agyroscope, or other suitable motion sensing technology to detect motionof the communication device 5800 in three-dimensional space. Theorientation sensor 5820 can utilize orientation sensing technology suchas a magnetometer to detect the orientation of the communication device5800 (north, south, west, and east, as well as combined orientations indegrees, minutes, or other suitable orientation metrics).

The communication device 5800 can use the transceiver 5802 to alsodetermine a proximity to a cellular, WiFi, Bluetooth, or other wirelessaccess points by sensing techniques such as utilizing a received signalstrength indicator (RSSI) and/or signal time of arrival (TOA) or time offlight (TOF) measurements. The controller 5806 can utilize computingtechnologies such as a microprocessor, a digital signal processor (DSP),programmable gate arrays, application specific integrated circuits,and/or a video processor with associated storage memory such as Flash,ROM, RAM, SRAM, DRAM or other storage technologies for executingcomputer instructions, controlling, and processing data supplied by theaforementioned components of the communication device 400.

Other components not shown in FIG. 58 can be used in one or moreembodiments of the subject disclosure. For instance, the communicationdevice 5800 can include a reset button (not shown). The reset button canbe used to reset the controller 5806 of the communication device 5800.In yet another embodiment, the communication device 5800 can alsoinclude a factory default setting button positioned, for example, belowa small hole in a housing assembly of the communication device 5800 toforce the communication device 5800 to re-establish factory settings. Inthis embodiment, a user can use a protruding object such as a pen orpaper clip tip to reach into the hole and depress the default settingbutton. The communication device 400 can also include a slot for addingor removing an identity module such as a Subscriber Identity Module(SIM) card. SIM cards can be used for identifying subscriber services,executing programs, storing subscriber data, and so forth.

The communication device 5800 as described herein can operate with moreor less of the circuit components shown in FIG. 58. These variantembodiments can be used in one or more embodiments of the subjectdisclosure.

It should be understood that devices described in the exemplaryembodiments can be in communication with each other via various wirelessand/or wired methodologies. The methodologies can be links that aredescribed as coupled, connected and so forth, which can includeunidirectional and/or bidirectional communication over wireless pathsand/or wired paths that utilize one or more of various protocols ormethodologies, where the coupling and/or connection can be direct (e.g.,no intervening processing device) and/or indirect (e.g., an intermediaryprocessing device such as a router).

FIG. 59 depicts an exemplary diagrammatic representation of a machine inthe form of a computer system 5900 within which a set of instructions,when executed, may cause the machine to perform any one or more of theembodiments described above. One or more instances of the machine canutilize the aforementioned adaptive antenna embodiments singly or in anycombination. In some embodiments, the machine may be connected (e.g.,using a network 5926) to other machines. In a networked deployment, themachine may operate in the capacity of a server or a client user machinein server-client user network environment, or as a peer machine in apeer-to-peer (or distributed) network environment.

The machine may comprise a server computer, a client user computer, apersonal computer (PC), a tablet PC, a smart phone, a laptop computer, adesktop computer, a control system, a network router, switch or bridge,or any machine capable of executing a set of instructions (sequential orotherwise) that specify actions to be taken by that machine. It will beunderstood that a communication device of the subject disclosureincludes broadly any electronic device that provides voice, video ordata communication. Further, while a single machine is illustrated, theterm “machine” shall also be taken to include any collection of machinesthat individually or jointly execute a set (or multiple sets) ofinstructions to perform any one or more of the methods discussed herein.

The computer system 5900 may include a processor (or controller) 5902(e.g., a central processing unit (CPU), a graphics processing unit (GPU,or both), a main memory 5904 and a static memory 5906, which communicatewith each other via a bus 5908. The computer system 5900 may furtherinclude a display unit 5910 (e.g., a liquid crystal display (LCD), aflat panel, or a solid state display. The computer system 5900 mayinclude an input device 5912 (e.g., a keyboard), a cursor control device5914 (e.g., a mouse), a disk drive unit 5916, a signal generation device5918 (e.g., a speaker or remote control) and a network interface device5920. In distributed environments, the embodiments described in thesubject disclosure can be adapted to utilize multiple display units 5910controlled by two or more computer systems 5900. In this configuration,presentations described by the subject disclosure may in part be shownin a first of the display units 5910, while the remaining portion ispresented in a second of the display units 5910.

The disk drive unit 5916 may include a tangible computer-readablestorage medium 5922 on which is stored one or more sets of instructions(e.g., software 5924) embodying any one or more of the methods orfunctions described herein, including those methods illustrated above.The instructions 5924 may also reside, completely or at least partially,within the main memory 5904, the static memory 5906, and/or within theprocessor 5902 during execution thereof by the computer system 5900. Themain memory 5904 and the processor 5902 also may constitute tangiblecomputer-readable storage media.

Dedicated hardware implementations including, but not limited to,application specific integrated circuits, programmable logic arrays andother hardware devices that can likewise be constructed to implement themethods described herein. Application specific integrated circuits andprogrammable logic array can use downloadable instructions for executingstate machines and/or circuit configurations to implement embodiments ofthe subject disclosure. Applications that may include the apparatus andsystems of various embodiments broadly include a variety of electronicand computer systems. Some embodiments implement functions in two ormore specific interconnected hardware modules or devices with relatedcontrol and data signals communicated between and through the modules,or as portions of an application-specific integrated circuit. Thus, theexample system is applicable to software, firmware, and hardwareimplementations.

In accordance with various embodiments of the subject disclosure, theoperations or methods described herein are intended for operation assoftware programs or instructions running on or executed by a computerprocessor or other computing device, and which may include other formsof instructions manifested as a state machine implemented with logiccomponents in an application specific integrated circuit or fieldprogrammable gate array. Furthermore, software implementations (e.g.,software programs, instructions, etc.) including, but not limited to,distributed processing or component/object distributed processing,parallel processing, or virtual machine processing can also beconstructed to implement the methods described herein. It is furthernoted that a computing device such as a processor, a controller, a statemachine or other suitable device for executing instructions to performoperations or methods may perform such operations directly or indirectlyby way of one or more intermediate devices directed by the computingdevice.

While the tangible computer-readable storage medium 5922 is shown in anexample embodiment to be a single medium, the term “tangiblecomputer-readable storage medium” should be taken to include a singlemedium or multiple media (e.g., a centralized or distributed database,and/or associated caches and servers) that store the one or more sets ofinstructions. The term “tangible computer-readable storage medium” shallalso be taken to include any non-transitory medium that is capable ofstoring or encoding a set of instructions for execution by the machineand that cause the machine to perform any one or more of the methods ofthe subject disclosure.

The term “tangible computer-readable storage medium” shall accordinglybe taken to include, but not be limited to: solid-state memories such asa memory card or other package that houses one or more read-only(non-volatile) memories, random access memories, or other re-writable(volatile) memories, a magneto-optical or optical medium such as a diskor tape, or other tangible media which can be used to store information.Accordingly, the disclosure is considered to include any one or more ofa tangible computer-readable storage medium, as listed herein andincluding art-recognized equivalents and successor media, in which thesoftware implementations herein are stored.

Although the present specification describes components and functionsimplemented in the embodiments with reference to particular standardsand protocols, the disclosure is not limited to such standards andprotocols. Each of the standards for Internet and other packet switchednetwork transmission (e.g., TCP/IP, UDP/IP, HTML, HTTP) representexamples of the state of the art. Such standards are from time-to-timesuperseded by faster or more efficient equivalents having essentiallythe same functions. Wireless standards for device detection (e.g.,RFID), short-range communications (e.g., Bluetooth, WiFi, Zigbee), andlong-range communications (e.g., WiMAX, GSM, CDMA, LTE) can be used bycomputer system 5900.

The illustrations of embodiments described herein are intended toprovide a general understanding of the structure of various embodiments,and they are not intended to serve as a complete description of all theelements and features of apparatus and systems that might make use ofthe structures described herein. Many other embodiments will be apparentto those of skill in the art upon reviewing the above description. Theexemplary embodiments can include combinations of features and/or stepsfrom multiple embodiments. Other embodiments may be utilized and derivedtherefrom, such that structural and logical substitutions and changesmay be made without departing from the scope of this disclosure. Figuresare also merely representational and may not be drawn to scale. Certainproportions thereof may be exaggerated, while others may be minimized.Accordingly, the specification and drawings are to be regarded in anillustrative rather than a restrictive sense.

Although specific embodiments have been illustrated and describedherein, it should be appreciated that any arrangement calculated toachieve the same purpose may be substituted for the specific embodimentsshown. This disclosure is intended to cover any and all adaptations orvariations of various embodiments. Combinations of the aboveembodiments, and other embodiments not specifically described herein,can be used in the subject disclosure.

The Abstract of the Disclosure is provided with the understanding thatit will not be used to interpret or limit the scope or meaning of theclaims. In addition, in the foregoing Detailed Description, it can beseen that various features are grouped together in a single embodimentfor the purpose of streamlining the disclosure. This method ofdisclosure is not to be interpreted as reflecting an intention that theclaimed embodiments require more features than are expressly recited ineach claim. Rather, as the following claims reflect, inventive subjectmatter lies in less than all features of a single disclosed embodiment.Thus the following claims are hereby incorporated into the DetailedDescription, with each claim standing on its own as a separately claimedsubject matter.

It is to be understood that although the disclosure has been describedabove in terms of particular embodiments, the foregoing embodiments areprovided as illustrative only, and do not limit or define the scope ofthe disclosure.

Various other embodiments, including but not limited to the following,are also within the scope of the claims. For example, the elements orcomponents of the various multimode antenna structures described hereinmay be further divided into additional components or joined together toform fewer components for performing the same functions. For example,the antenna elements and the connecting element or elements that arepart of a multimode antenna structure may be combined to form a singleradiating structure having multiple feed points operatively coupled to aplurality of antenna ports or feed points.

It is further noted that the low band and high band antennae structuresdescribed in the subject disclosure may be different or dissimilarantenna types, such as, for example, monopole, PIFA, loop, dielectric orother structures known in the art. It is also noted that the embodimentsdescribed herein may represent other sub-frequency ranges such as, forexample, low band, mid band, and high band. Accordingly, the antennastructures described herein may have differing antenna types, anddiffering frequency ranges.

Having described embodiments of the present disclosure, it should beapparent that modifications can be made without departing from thespirit and scope of the disclosure.

What is claimed is:
 1. A method, comprising: measuring, by a circuit, achange in reactance of an antenna; determining, by the circuit, afrequency offset of the antenna based on a change in an operatingfrequency of the antenna according to the change in reactance of theantenna; and adjusting, by the circuit, the operating frequency of theantenna to mitigate the frequency offset of the antenna.
 2. The methodof claim 1, wherein the circuit comprises a reactive sensor.
 3. Themethod of claim 2, wherein the reactive sensor comprises a capacitivesensor.
 4. The method of claim 1, wherein determining the change in theoperating frequency of the antenna comprises: determining a currentoperating frequency of the antenna; obtaining impedance characteristicsof the antenna according to the current operating frequency of theantenna; and determining the frequency offset according to the change inreactance measured and the impedance characteristics.
 5. The method ofclaim 4, wherein the obtaining of the impedance characteristics of theantenna comprises retrieving the impedance characteristics of theantenna from a look-up table.
 6. The method of claim 5, wherein thelook-up table comprises a plurality of impedance characteristics of theantenna measured at a plurality of operating frequencies of the antenna.7. The method of claim 1, wherein the determining the frequency offsetof the antenna further comprises obtaining an impedance of a matchingnetwork coupled to the antenna.
 8. The method of claim 1, wherein thechange in reactance of the antenna comprises a change in a capacitanceof the antenna.
 9. The method of claim 1, wherein the frequency of theantenna is adjusted by modifying an electrical length of the antenna.10. The method of claim 1, wherein the antenna comprises an aperturetuner to adjust a resonant frequency range of the antenna, and whereinthe adjusting of the operating frequency of the antenna is performed bysupplying a signal to the aperture tuner.
 11. The method of claim 10,wherein the aperture tuner comprises a variable reactive element toadjust the resonant frequency range of the antenna.
 12. The method ofclaim 10, wherein the aperture tuner comprises a switchable array ofreactive elements to adjust the resonant frequency range of the antenna.13. The method of claim 10, wherein the aperture tuner comprises one ofa variable capacitor, a variable inductor, or a combination thereof. 14.An antenna structure, comprising: a first antenna element; and a sensorcoupled to the first antenna element, wherein the sensor is coupled to acircuit that performs operations comprising: measuring from the sensor achange in a reactance of the antenna; obtaining impedancecharacteristics of the antenna; determining a change in an operatingfrequency of the antenna according to the change in reactance of theantenna and the impedance characteristics of the antenna; and adjustingthe operating frequency of the antenna to counteract the change in theoperating frequency of the antenna.
 15. The antenna structure of claim14, wherein the sensor measures a change in a capacitance of theantenna, a change in an inductance of the antenna, or a combinationthereof.
 16. The antenna structure of claim 14, wherein the obtaining ofthe impedance characteristics of the antenna comprises: determining acurrent operating frequency of the antenna; and obtaining from a look-uptable the impedance characteristics of the antenna according to thecurrent operating frequency of the antenna.
 17. The antenna structure ofclaim 14, wherein the operating frequency of the antenna is adjusted bymodifying an electrical length of the antenna.
 18. A communicationdevice, comprising: an antenna structure; a sensor; and a circuitcoupled to the sensor, wherein the circuit performs operationscomprising: measuring a change in a reactance of the antenna;determining a frequency offset of the antenna according to the change inreactance of the antenna; and adjusting an operating frequency of theantenna to reduce the frequency offset of the antenna.
 19. Thecommunication device of claim 18, wherein the sensor comprises areactance sensor.
 20. The communication device of claim 18, wherein theoperating frequency of the antenna is adjusted by modifying anelectrical length of the antenna.